1.1 Introduction

Electromagnetic Compatibility (EMC) is defined as the ability of a device, equipment or a system to function satisfactorily in its electromagnetic environment without introducing intolerable electromagnetic disturbance to anything in that environment. Any electronic equipment is both capable of emitting unintended signals (i.e., interference to other electronic equipment) and also itself being affected by spurious radiation from other electronic equipment (i.e., interference caused by other electronic equipment). Putting these electronics together without affecting each other is the challenge. Meeting the challenge is a combination of legislation, engineering and consideration for the needs of others.

1.2 EMI vs EMC

Electromagnetic interference (EMI) is a serious and increasing form of environmental pollution. Its effects range from extremely small annoyances due to crackles on broadcast reception, to potentially fatal accidents due to corruption of safety-critical control systems. Various forms of EMI may cause electrical and electronic malfunctions, can prevent the proper use of the radio frequency spectrum, can ignite flammable or other hazardous atmospheres, and may even have a direct effect on human tissue. As electronic systems penetrate more deeply into all aspects of society, so both the potential for interference effects and the potential for serious EMI-induced incidents will increase.

The threat of EMI is controlled by adopting the practices of electromagnetic compatibility (EMC). The term EMC has two complementary aspects:

• It describes the ability of electrical and electronic systems to operate without interfering with other systems
• It also describes the ability of such systems to operate as intended within a specified electromagnetic environment

Thus it is closely related to the environment within which the system operates. Effective EMC requires that the system is designed, manufactured and tested with regard to its predicted operational electromagnetic environmental (i.e., the totality of electromagnetic phenomena existing at its location). Although the term electromagnetic tends to suggest an emphasis on high frequency field-related phenomena, in practice the definition of EMC encompasses all frequencies and coupling paths, from DC through mains supply frequencies and microwaves.

1.3 Interference sources

There are two types of internal and external interference, viz., continuous and intermittent. Each type has its own cause.

Continuous interference

Continuous, or Continuous Wave (CW), interference arises where the source continuously emits at a given range of frequencies. This type is naturally divided into sub-categories according to frequency range, and as a whole is sometimes referred to as “DC to daylight”.

• Audio Frequency, from very low frequencies up to around 20kHz. Frequencies up to 100kHz may sometimes be classified as Audio. Sources include:
  • Mains hum from: power supply units, nearby power supply wiring, transmission lines and substations.
  • Audio processing equipment, such as audio power amplifiers and loudspeakers.
  • Demodulation of a high-frequency carrier wave such as an FM radio transmission.
• Radio Frequency Interference (RFI), from typically 20kHz to an upper limit which constantly increases as technology pushes it higher. Sources include:
  • Wireless and Radio Frequency Transmissions
  • Television and Radio Receivers
  • Industrial, scientific and medical equipment (ISM)
  • Digital processing circuitry such as microcontrollers
• Broadband noise may be spread across parts of either or both frequency ranges, with no particular frequency accentuated. Sources include:
  • Solar activity
  • Continuously operating spark gaps such as arc welders
  • CDMA (spread-spectrum) mobile telephony

Pulse or transient interference

An electromagnetic pulse (EMP), sometimes called a transient disturbance, arises where the source emits a short-duration pulse of energy. The energy is usually broadband by nature, although it often excites a relatively narrow-band damped sine wave response in the victim.

Sources divide broadly into isolated and repetitive events.

• Sources of isolated EMP events include:
  • Switching action of electrical circuitry, including inductive loads such as relays, solenoids, or electric motors.
  • Electrostatic discharge (ESD), as a result of two charged objects coming into close proximity or contact.
  • Lightning electromagnetic pulse (LEMP), although typically a short series of pulses.
  • Nuclear electromagnetic pulse (NEMP), as a result of a nuclear explosion. A variant of this is the high altitude EMP (HEMP) nuclear device, designed to create the pulse as its primary destructive effect.
  • Non-nuclear electromagnetic pulse (NNEMP) weapons.
  • Power line surges/pulses
• Sources of repetitive EMP events, sometimes as regular pulse trains, include:
  • Electric motors
  • Gasoline engine ignition systems
  • Continual switching actions of digital electronic circuitry.

It is also possible to categorise the different types of EMI by their bandwidth.

• Narrowband: Typically this form of EMI is likely to be a single carrier source – possibly generated by an oscillator of some form. Another form of narrowband EMI is the spurious signals caused by intermodulation and other forms of distortion in a transmitter such as a mobile phone of Wi-Fi router. These spurious signals will appear at different points in the spectrum and may cause interference to another user of the radio spectrum. As such these spurious signals must be kept within tight limits.
• Broadband: There are many forms of broadband noise which can be experienced. It can arise from a great variety of sources. Man-made broadband interference can arise from sources such as arc welders where a spark is continuously generated. Naturally occurring broadband noise can be experienced from the Sun – it can cause sun-outs for satellite television systems when the Sun appears behind the satellite and noise can mask the wanted satellite signal. Fortunately these episodes only last for a few minutes.

The most common causes of continuous interference are:

• 50/60 Hz Supply Power
• Electric Motor (Especially Commutator Type)
• High Power Radio Signals
• Switch Mode Power Supplies
• Microwave Ovens
• Ignition Circuits

Devices that cause constant noise emissions are usually easier to find than intermittent noise problems. This is because the noise doesn’t go away while the system is being looked at. The most common constant noise source is hum, caused by a 50/60 Hz supply power. Supply power is the most common noise component because it is an oscillating voltage, has high power and has a huge antenna system. Almost every system has some form of power filtering for 50/60 Hz supply power. This filtering can take the form of either trying to keep 50/60 Hz noise from getting into our device or leaving the device.

High power electric motors often create wideband noise. They can radiate noise into almost any equipment that is in close proximity to the motor. DC motors often have switch mode power supplies that cause high frequency noise through the common power ground. As motors ramp up and down the noise can vary in frequency and power. This wideband motor noise can then be transmitted back through the power supply lines or through a common earth ground.

Local radio, television stations, radar and ham radio stations can cause radio frequency noise. Military radar is the highest power radio system, but TV, AM and FM local radio stations are usually more common. These stations put out kilowatts of power and often are relatively close to industrial areas.

Switch mode power supplies are fast becoming the most common noise source of all. This is because they are so popular as a low voltage plug pack in home electronics. They create large amounts of harmonic frequencies. The power supplies develop low voltages from mains power by switching the high voltage on and off very quickly that creates lots of noise. The wires that connect the power supply to the device then transmit this noise. Switch mode power supplies are very popular because they are not frequency or voltage dependent and can be used in any country and on almost any device.

Microwave ovens radiate wideband noise by leakage through edges of the door or from the power supply wires. The oven can transmit hundreds of watts over short distances. While this is good for cooking the food, it also can create life-threatening situations for people with pacemakers. Ignition circuits in motorcycles, cars and other gasoline powered motors put out wideband noise created by tens of thousands of Volts. Most automobiles have some sort of suppression circuit, but if this fails it can cause havoc with it’s own electronics and nearby devices. Older motorcycles, lawn mowers and other simple engines often have little or no noise suppression and therefore emit large amounts of noise.

Devices that cause intermittent wideband noise are:

• Lightning
• Switching Relay Equipment
• Arc Welding
• Static

Intermittent noise components are often hard to find. There is an old saying in electronics… If it’s not broke, it can’t be fixed. And since the noise comes and goes the problem is usually only present part of the time.

Lightning can be the most damaging of the intermittent noise. A typical lightning strike can contain 20 to 40 thousand Amps and millions of Volts. In addition, the lightning strike transmits wideband noise that covers the whole frequency spectrum from DC to X-rays. This, in conjunction with the high current and voltage, makes it impossible to filter out lightning noise. The best method is to keep the lightning away from the circuit by using protection devices like shunts and suppressors.

Turning off relays usually causes relay switching noise. This noise is created by the magnetic field collapsing when the relay is turned off. This type of noise is common in industrial environments.

Arc welding is man-made lightning and has all the attributes of lightning such as high current, high voltage and wideband frequency noise. The advantage here is that it is often very intermittent and can be easily recognized.

Static is very hard to prove as a noise source as it is often invisible and very intermittent. Although is often man-made, it can be natural in origin. Equipment can be spiked by static build up in the air as well as from a person. Static noise is also very similar to lightning with all the same attributes except on a smaller scale.

1.4 Need for standards

Standards are required to control interference from the electronic devices, i.e., to make electronic devices less susceptible to interference. Various countries implemented their own standards, some of the standards are as mentioned below:

• IEC (International Electrotechnical Commission) – the IEC operates in close co-operation with the International Organization (ISO). It is composed of National Committees that are expected to be fully representative of all electrotechnical interests in their respective countries. Two IEC technical committees are devoted full time to EMC work is TC77, Electromagnetic compatibility between equipment including networks, and the International Special Committee on Radio Interference or CISPR.
• CENELEC (European Committee for Electrotechnical Standardization) and ETSI (European Telecommunications Standards Institute) – CENELEC is the European standards making body, which has been mandated by the Commission of the EC (European Commission) to produce EMC standards for the use with European EMC Directive. For Telecommunications equipment ETSI is the mandated standards body. ETSI generates standards for telecom network equipment that is not available to the subscriber, and for the radio communications equipment and broadcast transmitters.

Any manufacturer wanting to market his/her goods into a particular country has to comply with the standards followed by that country.

1.5 EMC – the issues

Figure 1.3 shows the EMC issues in a diagrammatic way. Any electronic device will emit radiation in the form of –

• Radiated RF electromagnetic disturbances from the device itself
• Radiated RF electromagnetic disturbances from its input and output connections
• Conducted EMD via its I/O connections or power lines

Any electronic device will be susceptible to EMD from –

• Stray radiation from other electronic equipment
• Stray radiation from the I/O of other electronic equipment
• Conducted disturbances via mains lines or I/O, or
• Mains voltage variations or waveshape

1.6 Electromagnetic disturbances

Any electromagnetic phenomenon may degrade the performance of the system. Some of the electromagnetic issues are as mentioned below:

Voltage fluctuations: short-term (sub-second) fluctuations with quite small amplitudes are annoyingly perceptible on electric lightning, though electronic power supply circuits comfortably ignore them. Generation of flicker by high power load switching is subject to regulatory control.

Waveform distortion: at source, the AC mains is generated as a pure sine wave but the reactive impedance of the distribution network together with the harmonic currents drawn by non-linear loads causes voltage distortion. Power converters and electronic power supplies are important contributors to non-linear loading. Harmonic distortion may actually be worse at points remote from the non-linear load because of resonance in the network components. Not only must non-linear harmonic currents be limited but also equipment should be capable of operating with up to 10% total harmonic distortion in the supply waveform.

Voltage variations: the distribution network has finite source impedance and varying loads will affect the terminal voltage. Not including voltage drops within the customer’s premises, an allowance of ±10%, on the nominal voltage will cover normal variations in the UK. The effect of the shift in nominal voltage from 240 V to 230 V, as required by CENELEC Harmonization Document HD 472 S1: 1988 and implemented in the UK by BS 7697: 1993, is that from 1^st January 1995 the UK nominal voltage is 230 V with a tolerance of +10%, –6%. After 1^st January 2003 the nominal voltage will be 230 V with a tolerance of ±10%, in the line with all other Member States.

Transients and surges: switching operations generate transients of a few hundred volts as a result of current interruption in an inductive circuit. These transients normally occur in bursts and have risetimes of no more than a few nsec, although the finite bandwidth of the distribution network will quickly attenuate all but local sources. High amplitude spikes in excess of 2 kV may be observed due to fault conditions. Even higher voltage surges due to lightning strikes occur – mostly on exposed overhead line distribution systems in rural areas.

Voltage interruptions: faults on power distribution systems cause almost 100% voltage drops but are cleared quickly and automatically by protection devices, and throughout the rest of the system the voltage immediately recovers. Most consumers therefore see a short voltage dip. The frequency of occurrence of such dips depends on location and seasonal factors.

• Supply voltage – Mains electricity suffers a variety of disturbing effects during its distribution. These may be caused by sources in the supply network or by other users, or by other loads within the same installation. A pure, uninterrupted supply would not be cost effective; the balance between the cost of the supply and its quality is determined by national regulatory requirements, tempered by the experience of the supply utilities. Typical disturbances are:
  • # Interruptions
  • # Dips
  • # Surges
  • # Waveform distortion
  • # Fluctuations
  • # Voltage imbalance
  • # Frequency variations
  • # DC in AC networks
  • # Power-line carrier signaling
• Transient overvoltages – transient overvoltages on supply, signal and control lines that may occur because of lightning, switching or ESD (electrostatic discharge) can degrade the system performance. Take an example of an EMC phenomenon caused because of an ESD transient. An ESD transient can corrupt the operation of a microprocessor or clocked circuit just as a transient coupled into the supply or a signal port.
• Radio frequency fields – radio frequency fields, pulsed (radar), modulated and continuous, coupled directly into the equipment or onto its connected cables, may also degrade the system performance to a great extent.
• NEMP (nuclear electromagnetic pulse) – this is the effect of a high altitude nuclear explosion, which generates a sub-nanosecond nuclear electromagnetic pulse (NEMP) that is disruptive over an area of hundreds of square kilometers.
• LF magnetic or electric fields

1.7 EMC testing categories

Figure 1.4 shows the diagrammatic view of the EMC testing categories. The four categories are as follows:

• Conducted Emission (CE)
• Conducted Susceptibility (CS)
• Radiated Emission (RE)
• Radiated Susceptibility (RS)

All electronic devices are susceptible to electromagnetic disturbances, if it is not the case, then there would be no EMC issue. However, none of the electronic devices are completely immune and so the EMC problem still exists. Electromagnetic Susceptibility can be defined as the inability of a device, equipment or a system to perform without degradation in the presence of an electromagnetic disturbance. On the other hand, immunity to electromagnetic disturbances can be defined as the ability of a device, equipment or a system to perform without degradation in the presence of an electromagnetic disturbance.

1.8 The compatibility gap

The increasing susceptibility of electronic equipment to electromagnetic influence is being paralleled by an increasing pollution of the electromagnetic environment. Susceptibility is a function of the adoption of VLSI technology in the form of microprocessors, both to achieve new tasks and for those that were previously tackled by electromechanical or analog means, and the accompanying reduction in the energy required of potentially disturbing factors. It is also a function of the increased penetration of radio communications, and the greater opportunities for interference to radio reception that result from the co-location of unintentional emitters and radio receivers.

At the same time more radio communications means more transmitters and an increase in the average RF field strengths to which equipment is exposed. Also, the proliferation of digital electronics means an increase in low-level emissions that affect radio reception – a phenomenon that has been aptly described as a form of electromagnetic smog. Only legislation to limit the effects of interaction can solve the problem.

These concepts can be graphically presented in the form of a narrowing electromagnetic compatibility gap, as shown in Figure 1.5. This gap is of course conceptual rather than absolute, and the phenomena defined as emissions and those defined as immunity do not mutually interact except in rare cases. But the maintenance of some artificially-defined gap between equipment immunity and radio transmissions on the other hand, and equipment emissions and radio reception on the other, is the purpose of the application of EMC standards, and is one result of the enforcement of the EMC directive.

1.9 Emission, immunity and compatibility

Relation between emission, immunity and compatibility is as shown by graphical representation in Figure 1.6. As shown in the Figure, there are 3 different levels:

• The lowermost level is the Emission level; the maximum level in emission level is the emission limit. The emissions from the devices should be within this level and it should not exceed the emission limit.
• The uppermost level is the Immunity level; the minimum level in the immunity level is the immunity limit. The immunity of the devices should be within this level and it should not be less than the immunity limit.
• The middle level is the Compatibility level; the difference between the Immunity limit and the Emission limit is the Compatibility Margin. Greater the compatibility margin, lesser the chances of EMC problems.

For proper functioning of a system, the devices should be compatible (with respect to EM environment) and hence the term Compatibility.

1.10 Causes and consequences of EMI

The consequences of EMI can be classified in different categories depending on its criticality. Some of the causes of EMI that results into these consequences are mentioned below:

• Malfunction of a safety-critical item of machinery
• Erratic operation of moving equipment
• A safety device to ignore a signal
• An operation to stop for no apparent reason
• Not carry out its intended function, but not cause any havoc as a result

The malfunction will fall into the below mentioned classifications depending on the type of damage or loss occurred.

• Catastrophic – death, major injuries, downstream consequences
• Critical – minor injuries, extensive damage
• Major – minor permanent damage
• Minor – temporary performance loss
• Inconsequential – loss of performance within tolerance, no human intervention needed

1.11 Levels of compliance and EMC engineering application

1.11.1 Levels of compliance

There are different levels of EMC engineering. Figure 1.7 shows the compliance structure showing different levels at which designers and companies operate.

The four levels of compliance are as follows:

• Self-compatibility
• Minimum engineering/in-house standard
• Contractual compliance
• Special requirements

The above is useful to appreciate the change in engineering strategy – say a company was just struggling to make things works, but must now it must comply with specifications.

1.11.2 Levels of EMC engineering application

This course aims to equip with tools to design for and reduce EMI. Three main areas of application will be dealt with:

• PCBs
• Circuits/filters
• Screening

Proper grounding is underlying to all of the above areas and is also dealt with. Figure 1.8 shows the different areas.

PCBs can be seen as the inner or primary line of defence. The circuits on PCBs are where EMI problems eventually start and end. Proper PCB layout is the most subtle and cost effective way of influencing EMI. It controls EMI coupling right at the source or receptor – the circuit.

Filters and special circuits are used around the inner PCB as a secondary control measure or line of defence. When a PCB layout only does not eliminate unwanted interference, extra circuits and filters are added. Extra circuits imply more real estate and costs. Protection devices and circuits fall into this category.

Shielding is the tertiary or outer line of defence. This includes cables, screens and enclosures. If both of the above areas of application do not suffice, shielding is needed. The least cost effective solution and sometimes a last brute force attempt at compliance.

The fourth area of application (although not a level) is grounding. It determines the effectiveness and way all three levels interact. Grounding is applicable to all three of the above areas. Grounding on PCB level between different types of circuits is crucial. Filters/protection do not work properly if poorly grounded. Grounding of cable screens and enclosures has a primary influence on its shielding effectiveness.

2.1 Introduction

Electromagnetic interference (EMI) is caused in the presence of two parties, viz., a source and a victim. The source of interference is the aggressive circuits. High speed, noisy or dirty circuits come under the category of aggressive circuits. These are the noise generating circuits that also cause interference. The victims are sensitive circuits that are susceptible to interference. Sensitive, clean and quiet circuits come under the category of victim circuits. Any electronic device has the potential to radiate or conduct emission, and at the same time, they themselves are susceptible to it. In some cases, a victim itself acts as a source of interference. It is very important to understand which of the circuits (or modules) come under the category of ‘source’ and/or ‘victim’. Understanding such a mechanism helps in solving EMC problems.

2.2 Coupling paths: sources and victims

Figure 2.1 shows the potential interference routes that exist from one to the other when source and victim are put together. When the systems are built, it is necessary to know the emission signature and susceptibility of the component (equipment), along with problems that are likely to be experienced with close coupling. Following the published emission and susceptibility standards does not ensure freedom from system EMC problems. Standards are written for protecting a particular service, in the case of emissions, these are radio broadcast and telecommunications – they have to assume a minimum separation distance between source and victim. Most electronic devices consist of elements that are capable of antenna-like behaviour (i.e., they tend to radiate) such as cables, PCB tracks, internal wiring and mechanical structures. These elements can unintentionally transfer energy in the form of either electric, magnetic or electromagnetic fields that couple with the circuits. In practical situations, intra-system and external coupling between equipment is generally modified by the presence of screening and dielectric materials, and by the layout and proximity of interfering and victim equipment and especially their respective cables. Ground or screening planes tend to enhance interfering signal by reflection or attenuate it by absorption. Cable-to-cable coupling can be either inductive or capacitive and it depends on orientation, length and proximity. In most practical situations, dielectric materials may also reduce the field by absorption, though this is negligible compared to the effects of conductors.

2.2.1 Universal interference model

Figure 2.2 shows the working of EMI in the form of Universal Interference Model. It shows that there is always a noise source (i.e., culprit), receptor (i.e., victim) and coupling path (i.e., interference path).

The solution for reducing interference can be at the source, receptor or coupling path. The following are the means the reduce interference:

• Remove or reduce noise at source – low noise design, shielding, decoupling, etc.
• Improve victim immunity – high immunity design, shielding, filtering, decoupling, etc.
• Remove or attenuate coupling path – isolation, re-routing, shielding, spacing, filtering, etc.

2.3 Coupling mechanisms

There are many ways in which the electromagnetic interference can be coupled from the source to the receiver. Understanding which coupling method brings the interference to the receiver is key to being able to address the problem.

• Radiated: This type of EMI coupling is probably the most obvious. It is the type of EMI coupling that is normally experienced when the source and victim are separated by a large distance – typically more than a wavelength. The source radiates a signal which may be wanted or unwanted, and the victim receives it in a way that disrupts its performance.
• Conducted : Conducted emissions occur as the name implies when there is a conduction route along which the signals can travel. This may be along power cables or other interconnection cabling. The conduction may be in one of two modes:
  • Common mode: This type of EMI coupling occurs when the noise appears in the same phase on the two conductors, e.g. out and return for signals, or +ve and -ve for power cables.
  • Differential mode: This occurs when the noise is out of phase on the two conductors.
• The filtering techniques required will vary according to the type of EMI coupling experienced. For common mode lines are filtered together. For differential mode they may be filtered together.
• Inductive coupling: What is normally termed inductive coupling can be one of two forms, namely capacitive coupling and magnetic induction.
• Capacitive coupling : This occurs when a changing voltage from the source capacitively transfers a charge to the victim circuitry.
• Magnetic coupling: This type of EMI coupling exists when a varying magnetic field exists between the source and victim – typically two conductors may run close together (less than λ apart). This induces a current in the victim circuitry, thereby transferring the signal from source to victim.

By determining the form of coupling that exists and the way in which it is reaching the victim, it may prove to be that the most effective method of reducing the EMI is by putting measures in place to reduce the coupling and reduce the level of interference to an acceptable level.

Electromagnetic interference, EMI is present in all areas of electronics. By understanding the source, the coupling methods and the susceptibility of the victim, the level of interference can be reduced to a level where the EMI causes no undue degradation in performance.

Any EMI problem will involve any or a combination of the below mentioned coupling mechanism.

• Common impedance (Conducted)
• Capacitive (Electrostatic)
• Inductive (Magnetic)
• Electromagnetic (Radiated)

2.3.1 Common impedance (conducted)

From Figure 2.3, when an interference source (i.e., output of system A) shares a ground connection with a victim (i.e., input of system B) then any current due to A’s output flowing through the common impedance section (denoted by inductance L in Figure 2.3) develops a voltage in series with B’s input. The common impedance need not be more than a length of wire or pcb track, but high frequency or high di/dt components in the output will couple more efficiently because of the inductive nature of the impedance. The voltage developed across an inductor as a result of current flow through it is given by the equation 1 below:

V = –L×dI_L/dt  …………….(1)

where, L is the self inductance in H (Henries).

The output and input may be part of the same system; in this case there is a spurious feedback path through the common impedance that can cause oscillation. The solution as shown in Figure 2.3 is to simply re-route the connections so that there is no common current path, and hence no common impedance between the two circuits. The only penalty for doing this is the requirement for extra wiring or track to define the separate circuits. This applies to any circuit that may include common impedance (such as power rail connections). Grounds are most usual source of common impedance because the ground connection (generally not shown on circuit diagrams) is often taken for granted.

2.3.2 Inductive (magnetic)

AC current flowing in a conductor creates a magnetic field that will couple with a nearby conductor and induce a voltage in it as shown in Figure 2.4. The voltage induced in the victim conductor is now given by equation 2 below:

Vn = –M×dI_L/dt  …………….(2)

where, M is mutual inductance in H (Henries).

Notice the similarity between equation 2 and equation 1; M depends on the areas of the source and victim current loops, their orientation and separation distance, and the presence of any magnetic screening. Whether or not there is a direct connection between the two circuits, the coupling remains unaffected; the induced voltage would be the same even if the circuits were isolated or connected to ground.

2.3.3 Capacitive (electrostatic)

Changing voltage on one conductor creates an electric field that may couple with a nearby conductor and induce a voltage on it, as shown in Figure 2.5. The voltage induced on the victim conductor is given by equation 3 below:

V_n = C_C × dV_L/dt × Z_in//R_S   …………….(3)

where, C_C is the coupling capacitance,

Z_in//R_S is the impedance to ground of the victim circuit.

This assumes that the impedance of the coupling capacitance is much higher than that of the circuit impedances. The noise is injected as if from a current source with a value of C_C . dV_L/dt. The value of C_C is a function of the distance between the conductors, their effective areas and the presence of any electric screening material.

2.3.4 Real world coupling

A combination of coupling mechanisms exists in the real world. One type of coupling can dominate, but sometimes a combination is present as shown in Figure 2.6.

2.3.5 Mutual C and L vs Lead Spacing

Both mutual capacitance and inductance are affected by the physical separation of the source and victim conductors. Figure 2.7 is the graph showing the effects of the spacing between conductors on their ability to couple from one circuit to another.

2.4 Coupling via the supply network

Propagation of interference from a source to a victim can take place via the mains distribution network to which both are connected. This is not well characterized at high frequencies, since the electrical loads connected can present virtually any RF impedance at their point of connection. From Figure 2.8 we can see that the RF impedance presented by the mains can approximate a network of 50 Ω in parallel with 50 μH on the average. For short distances (such as that between adjacent outlets on the same ring) coupling via the mains connection of two items of equipment can be represented by the Figure 2.8.

Over longer distances, power cables are fairly low loss transmission lines having characteristic impedance of about 150-200 Ω up to about 10 MHz. However, in any local power distribution system the disturbances and discontinuities introduced by load connections, cable junctions and distribution components dominates the RF transmission characteristic, in turn these all tends to increase the attenuation.

2.5 Electromagnetic fields

Two conductors at different potentials generate an electric field (E field) between them. The field is proportional to the applied voltage divided by the distance between the conductors and is measured in volts per meter (V/m).

A conductor carrying a current generate magnetic field (H field) around it. The field is proportional to the current divided by the distance from the conductor and is measured in amps per meter (A/m).

When an alternating voltage generates an alternating current through a network of conductors (this description applies to any electronic circuit), an electromagnetic (EM) wave is generated that propagates as a combination of E and H fields. The speed of propagation of EM wave is determined by the medium (in free space it is equal to the speed of light, i.e., 3×10⁸ m/s). Near to the radiating source, the geometry and strength of the fields depend on the characteristics of the source. A circuit node carrying a significant dv/dt (rate of change of voltage) will generate mostly an electric field; a conductor carrying a significant di/dt (rate of change of current) will generate mostly a magnetic field. The structure of these generated fields will be determined by the physical layout of the source conductors, as well as by other conductors, dielectrics and permeable materials in the vicinity. Further away from the source, the complex three-dimensional field structure decays and only the components that are orthogonal to each other and to the direction of propagation remain. Figure 2.9 demonstrates these ideas graphically.

2.6 Rayleigh/Maxwell near/far fields

The ratio of the electric to magnetic field strengths (i.e., E/H) is called the wave impedance (Figure 2.10). The wave impedance is a key parameter of any given wave as it determines the efficiency of coupling with another conducting structure, and also the effectiveness of any conducting screen that is used to block it. Radiated emissions can be divided into near field and far field.

In the far field, i.e., for d > λ/2π, the wave is known as a plane wave and the E and H fields decline with distance at the same rate. Therefore its impedance is constant, and is equal to the impedance of free space given by equation 4:

Z_o = √μ_o/ɛ_o = 120π = 377 Ω   …………….(4)

where, μ_o = 4π . 10^-7 H/m, permeability of free space

ɛ_o = 8.85 . 10^-12 F/m, permittivity of free space

In the near field, i.e., for d < λ/2π, the wave impedance is determined by the characteristics of the source. Separate electric and magnetic field exist and which one will predominate depends on the source impedance. A low current, high voltage radiator such as a rod will mainly generate an electric field of high impedance, while a high current, low voltage radiator such as a loop will mainly generate a magnetic field of low impedance. As a special case, if the radiating structure happens to have impedance around 377Ω, then depending on the geometry, a plane wave can be generated in the near field.

The region around λ/2π (approximately one sixth of a wavelength) is the transition region between near and far fields. It indicates the region within which the field structure changes from complex to simple. Plane waves are always assumed to be in the far field, while if you are looking at the near field it is necessary to consider individual electric or magnetic fields separately.

2.7 Coupling modes

The fundamental to an understanding of EMC are the concepts of differential mode, common mode and antenna mode radiated field coupling. They apply to coupling of both emissions and interference.

2.7.2 Common mode

The cable also carries currents in common mode, i.e., all flowing in the same direction of each wire as shown in Figure 2.11. These currents very often have nothing at all to do with the signal currents. They may be induced by an external field coupling to the loop formed by the cable, the ground plane and the various impedances connecting the equipment to ground, then may cause internal differential currents (to which the equipment is susceptible). Unconventionally, they may be generated by internal noise voltages between the ground reference point and the cable connection, and are responsible for radiated emissions. The stray capacitances and inductances associated with the wiring and enclosure of each unit are an integral part of the common mode coupling circuit and play a major role in determining the amplitude and spectral distribution of the common mode currents. These stray impedances are incidental rather than designed in to the equipment. They don’t appear on any circuit diagram and are difficult to control or predict.

2.8 Radiated emissions from PCB (differential mode)

In most equipment, the primary sources are currents flowing in circuits such as clocks, oscillators, etc that are mounted on the PCB, some of the energy that is directly radiated from the PCB is modeled as a small loop antenna carrying the interference current. Figure 2.12 describes this situation. A small loop is that which has dimension smaller than a quarter wavelength (λ/4) at the frequency of interest (typical example – 1meter at 75 MHz). When the dimension of the loop approach λ/4 the currents at different points on the loop appear to be out of phase at a distance, so that the effect is to reduce the field strength at any given point. The electric field strength varies with the square of the frequency, and is directly proportional to the signal current and loop area.

E = 263 × 10^-12 × (f² × A × I_s) V/m   …………………(6)

where, A is loop area in cm²,

f (MHz) is the frequency of I_s the source current in mA.

From equation 6, it is clear that the field strength increases with the loop area. The loop area is the path traced by the signal current along with the return path. For the field strength to be as small as possible, the loop area should be small, so one of the rules in PCB design and layout is to keep the area covered by the signal current as small as possible (keep the loop areas small). Keep the signal return path as simple and clear as possible. Avoid puzzling and persistent signal paths as it may cover a large unintended area and that in turn results in increased field strength.

The loop antenna not only acts as a source of emission of unwanted noise but also can receive unwanted noise and in turn is the victim of interference. This is one of the main reason of using gridded grounds and ground planes, which ensures a defined return path for signal currents and avoids any unintentional return paths.

2.9 Cable radiated emissions (common mode)

Differential mode radiations from small loops on PCBs are not the only contributor to radiated emission; common mode currents flowing on the PCB and on attached cables can contribute much more in comparison. The differential mode currents that are governed by Kirchoff’s current law can be easily predicted, in contrast with the common mode currents on the PCB are not easy to predict. Figure 2.13 shows the return path for common mode currents is via stray capacitance (displacement current) to the other nearby objects.

The full prediction would therefore have to take into account the detailed structure (mechanical) of the PCB and its case, its proximity to ground and to other equipment. The interference current generated in common mode from ground noise developed across the PCB or elsewhere in the equipment and may flow along the conductors or along the shield of the shielded cable. Here even if the cables are longer than λ/20, it will act as an antenna. The common mode noise generally couple through parasitic capacitances.

The equation of field strength in case of common mode is as follows:

E = 1.26 × 10^-4 × (f × L ×I_CM) V/m   ……………..(7)

where, L is the cable length in meters,

I_CM is the common mode current at f MHz in mA flowing in the cable.

Here if the length of the cable increases, the field strength increases proportionally. In PCB design and layout, steps should be taken to reduce the common mode currents (minimize common mode currents). Partitioning, filtering, grounding and using planes in PCB design can achieve this.

2.10 Coupling paths (conducted emissions)

As in Figure 2.14, where there are high switch-mode frequencies, parasitic capacitance to ground plays a major part. This allows large common-mode voltages to develop relative to ground. In addition, differential mode signals can appear on the supply line or the signal cable as a result of SMPS noise getting through to the signal cable from the supply lines, or directly onto the Live and Neutral from the switching oscillator.

Figure 2.14 shows a typical product with a switched mode supply that gives an idea of the various paths these emissions can take. Differential mode current I_DM generated at the input of the switching supply is measured as an interference voltage across the load impendence of each with respect to earth at the measurement point. Higher frequency switching noise components V_Nsupply are coupled through C_C, the coupling capacitance between primary and secondary of the isolating transformer, to appear between L/N and E on the mains cable, and C_S to appear with respect to the ground plane. Circuit ground noise (digital noise and clock harmonics) is referenced to ground by C_S and coupled out via signal cables as I_CMsig (current through the signal cable) or via the safety earth as I_CME.

The problem in real situation is that all these mechanism are operating simultaneously, and the stray capacitances C_S are widely distributed and unpredictable, depending heavily on proximity to other objects if the case is unscreened. A partially screened enclosure may actually worsen the coupling because of its higher capacitance to the environment.

2.11 Susceptibility to radiated field coupling

Coupling can take place either directly with the internal circuitry and wiring in differential mode or with the cables to induce a common mode current as shown in Figure 2.15. Since wiring lengths of a few inches approach resonance at these frequencies, coupling with internal wiring and PCB tracks is most efficient at frequencies above a few hundred MHz.

2.12 Transient sources

Transients and spikes are different from continuously generated EMI. The below are the likely sources:

• Electro static discharge (ESD)
• Lightning
• Switching
• Power

2.12.1 Transients – Lightning, Switching and ESD

Virtually all the transient waveforms can be classified as shown in Figure 2.16.

There is some variation in from where the second time period, i.e., T2 start. In Figure 2.16 it is shown as being from almost the start of the rise of the wave. However, in most cases, both of these times are specified as being ±30%.

2.12.2 Transient frequency

The Figure 2.17 shows the result of a study carried out on the mains supply and the telecom lines to record the number of amplitudes of the transient voltages. These figures obviously depend on the lightning strike density in the various parts of the world, and the degree of heavy load switching in the vicinity of the particular site. Particularly with lightning (but also with switching surges) the mains connection density plays a part.

2.13 Automotive transients

Fast transients can be coupled (usually capacitively) onto signal cables in common mode, especially if the cable passes close to or is routed alongside an impulsive interference source. Although such transients are generally lower in amplitude as compared to the mains-borne ones, they are coupled directly into the I/O ports of the circuit and will therefore flow in the circuit ground traces, unless the cable is properly screened and terminated or the interface is properly filtered.

Other sources of conducted transients are the automotive 12 V supply. The automotive environment regularly experiences transients that are many times nominal supply range. The most serious automotive transients as shown in Figure 2.18 are as follows:

• The load dump that occurs when the alternator load is suddenly disconnected during heavy charging
• Switching of inductive loads such as motors and solenoids, and
• Alternator field decay that generates a negative voltage spike when the ignition switch is turned OFF.

2.14 Supply voltage phenomenon

The supply voltage can exhibit a variety of disturbances as shown in Figure 2.19.

The ITIC (Information Technology Industries Council) curve shown in Figure 2.20, demonstrates the fluctuation levels and time periods that are likely to upset a PC. From this it is obvious that short-duration voltage variations can definitely affect electronic equipment.

It is just as necessary to ensure that mains-powered equipment doesn’t introduce any of these mentioned phenomena:

• Voltage dips (short-duration reductions in voltage)
• Interruptions (complete absence of power for longer than 3 seconds)
• Harmonics
• Unbalance (voltage differences between phases)
• Flicker (rapid voltage variations which are annoying as they affect incandescent lighting)
• Transients

3.1 Introduction

One of the most important aspects in EMC is to understand the difference between the possible modes of coupling. The basis of this differentiation is the idea that two separate circuit paths can coexist in the same set of conductors. The two coupling modes are as mentioned below:

Differential mode – differential simply means the difference between things of the same kind. Differential mode is the normal voltage and current between the signal and its return lines (or for that matter, it can be the positive and negative points in a circuit). Differential (in this case) means the difference between the two lines.

Common mode – common mode comes into the picture when there is another ground (reference) plane with respect to which voltages can exist and currents can flow. Common (here) means common between the two lines and a reference common to them. In common mode the two lines are seen as one.

EMI reduction has a lot to do with solving common mode problems. Common mode paths and voltages are more difficult to understand and visualize in comparison to differential mode. The latter involves current flowing as a normal circuit operation i.e., in loop. However, Common mode does not involve current flowing as per normal circuit operation as it involves multiple coupling paths and parasitic elements.

3.2 DM/CM conversion

Although it is said that common mode currents might be unrelated to the intended signal currents, there may also be a component of common mode current that is due to the signal current. Conversion occurs when the two signal conductors present different impedances to their environment, which is represented by the external ground. These impedances are dominated at RF by stray inductance and capacitance that relate to physical layout and are only under the circuit designer’s control (if and only if that person is also responsible for physical layout).

In Figure 3.1 the differential mode current I_DM produces the desired signal voltage across load R_L. The common mode current I_CM does not flow through load R_L directly but through impedances Z_A and Z_B, and returns back through the external ground. Z_A and Z_B are not circuit components but distributed stray impedances (typically but not always capacitive) and are determined by factors such as the surface area of PCB tracks and components and their proximity to chassis metalwork and other parts of the equipment. If Z_A = Z_B then no voltage is developed across R_L by the common mode currents I_CM. But any inequality (i.e., Z_A ≠ Z_B) results in such a voltage, proportional to the differences in impedance as given by equation 1:

V_load(CM) = I_CM.Z_A – I_CM.Z_B = I_CM. (Z_A – Z_B)  …………………..(1)

For this reason, circuits that carry high-frequency signals (such as wideband data or video) or which could be susceptible to RF are best designed in such a way that the stray impedances of each conductor are balanced as near as possible. Alternatively, a common-mode choke is used that swamps the imbalance of the strays and reduces the magnitude of common mode current I_CM. The increasing popularity of wideband data transmission through unscreened cables within and between buildings has sharpened the problem of interference radiated from these cables. The balance of the circuit at either end of the cable, the balance of the cable itself as it passes near to other conducting structures in its environment (vicinity), is a prime factor. This is largely determined by the quality of the cable construction, and has resulted in a cable parameter known as longitudinal conversion loss (LCL). The LCL of a balanced cable system is a measure of the mode conversion shown by the system, i.e. the degree to which an inadequately balanced termination and unwanted differential signal will develop when excited by a common mode signal.

3.3 Common mode rejection ratio (CMRR)

CMRR is a measure of how well a system can see the differential signal in the presence of common mode (CM) noise. Figure 3.2 shows such an arrangement to measure CMRR. In this case, the common mode voltage is applied to both the positive as well as the negative terminal of Opamp (Operational Amplifier), while differential mode voltage is applied separately to the positive and negative terminals of the Opamp. K is the gain provided by the Opamp. The voltage at the output of the Opamp due to a common mode signal is the average of voltages V_A and V_B. Therefore, it can be said that the output due to a common mode signal depends on the individual signal levels at point A and B. On the other hand, voltage at the output of an Opamp due to a differential mode signal is the difference between the voltages V_A and V_B. Therefore, it can be said that the output due to a differential mode signal does not depend on the individual signal levels at point A and B. CMRR can be represented by the equation 2 given below:

CMRR = 20log₁₀(V_CM/V_o)  …………………..(2)

where, V_CM = (V_A +V_B)/2 and

V_o = V_A – V_B

3.4 Units in EMC engineering

There is a wide range of units used right from DC (i.e., 0 Hz) to GHz using logarithmic scales for frequency and μA to kA, μV to MV, etc using decibels (dB) for magnitude. Decibel as a ratio is used to describe relative levels and relative power. A relative base unit such as 1 μV/m is used while describing levels.

X dBμV/m = 20log₁₀(X μV/m / 1 μV/m)  …………………..(3)

Examples of Decibel as a ratio are as follows:

0.1 μV/m = –20 dBμV/m

1 μV/m = 0 dBμV/m

3 μV/m = 10 dBμV/m

10 μV/m = 20 dBμV/m

100 μV/m = 40 dBμV/m

3.4.1 The decibel: power

Figure 3.3 shows V_in as the input voltage (P_in = corresponding input power) and V_out as the output voltage (P_out = corresponding output power). 0 dB gain means unity gain as shown in the Figure 3.3. It means that the input power is the same as the output power. Similarly, 3 dB gain doubles the power while, –3 dB gain halves the power and therefore 3 dB points are called half power point.

Power ratio and voltage ratio are given by the equations 4 and 5 respectively:

Power ratio = 10log₁₀(P_out/P_in)  …………………..(4)

Voltage ratio = 20log₁₀(V_out/V_in)  …………………..(5)

3.4.2 The decibel: signal to noise (S/N) ratio

‘Signal to noise ratio’, the name itself is self-explanatory; it means the ratio of signal level to the noise level. As shown in Figure 3.4, the signal level at the input is 5 V while the noise level is 0.05 V; therefore the output signal level is 5.05 V i.e., the sum of input signal level and the noise level attached to it. A signal to noise ratio in terms of dB can be represented by equation 6:

S/N = 20log₁₀(V_in/V_noise)  …………………..(6)

From, the example taken in Figure 3.4,

S/N = 20log₁₀(5/0.05) = 20log₁₀(100) = 40dB

The ratio should be as high as possible, i.e., ideally it should tend to infinity.

3.4.3 dBm

One of the most common units expressed in decibels is dB(mW) or dB relative to 1 milliwatt. This is almost always written in the abbreviated form, dBm (i.e. without the “W” and without the parentheses). Many oscilloscopes and spectrum analyzers optionally display voltage amplitudes in dBm. Since dBm is a unit of power, we must know the impedance of the measurement in order to convert dBm to volts. For example, a voltage expressed as 0 dBm on a 50-ohm spectrum analyzer is,

Example 1-3: Specifying voltages in dBm

Specify the following voltages in dBm assuming they were measured with a 50-ohm oscilloscope:

3.5 Spectrum usage and created Interference

Figure 3.5 shows the Electromagnetic spectrum that currently exists ranging from almost 0 Hz (i.e., DC) to around 1 GHz and includes AC supplies, AM, FM, cell-phones, etc. Any equipment required to stay immune should not cause interference within the same band of frequency or other bands of frequencies.

In reality, it is seen that virtually all electronic equipments potentially cause interference in almost all those areas where it is not acceptable. From an EMC point of view it is essential to know exactly where in the frequency spectrum a culprit exists i.e., causes interference and at what frequencies a victim is sensitive i.e., affected by interference caused by others or by itself. Figure 3.6 shows the operating frequencies of an AC supply, switch-mode power supplies and a microprocessor clock along with their harmonics. Due to 50/60 Hz rectifiers, thyristor switches, load switching etc., utility grid interference usually occurs in the Hz to kHz region. On the other hand, switch-mode power supplies operate/switch in the kHz region with harmonics extending into the MHz region.

3.6 Fourier analysis

Fourier analysis is used to switch between a time domain and a frequency domain. An oscilloscope is used for observing (or viewing) the signals in the time domain, similarly a spectrum analyzer is used for observing (or viewing) the signals in a frequency domain. If a sine wave of time period T is seen on the oscilloscope screen in the time domain, then the corresponding view in the frequency domain on a spectrum analyzer will be a single line as shown in Figure 3.7. The frequency at which this single line will appear on a spectrum analyzer will be F_o, where F_o = 1/T.

3.6.1 Frequency Spectrum of a Pulse

Figure 3.8 shows the frequency spectrum of a single pulse. A single pulse consists of a spectrum of frequencies. There are no distinct frequency components, but rather a smudge of components of variable amplitude. The critical parameter is F₂, the effective bandwidth of the pulse; after F₂ the spectrum decays quickly.

Figure 3.9 shows the frequency spectrum of a pulsed periodic signal. A pulsed periodic signal consists of a spectrum of discrete frequencies. The critical parameter is F₂ as it is the effective bandwidth of the signal. After F₂ the spectrum decays quickly.

3.7 Choice of logic family

The table shown in Figure 3.10 shows the logic family along with the corresponding rise time and bandwidth. The bandwidths for different logic families are calculated using the 1/(πt_r) formula. It shows that fast logic can generate a very large noise spectrum.

The damage (as far as emissions are concerned) is done by switching edges having fast rise times (note that this is not the same as propagation delay and is hardly specified in data sheets; where it is a maximum figure). Using the slowest risetime compatible with reliable operation minimizes the amplitude of the higher order harmonics where radiation is more efficient.

The advice is to use the slowest logic family that will do the job; don’t use the fast logic family. Where parts of the circuit operate at high speed, use fast logic only for those parts and keep the clock signals local. This preference for slow logic is unfortunately against the demands of software engineers (of greater processing speeds).

3.8 Lightning and ESD bandwidth

Although lightning and ESD are both pulsed noise, they differ in their treatment and threats. The following are the points that make them two distinct threats:

• The first and foremost is their different bandwidths, lightning is primarily a conductive immunity concern and ESD a radiated immunity concern.
• Lightning is typically a microsecond event; on the other hand, ESD is a nanosecond event.
• The typical bandwidth for lightning is around 300 kHz while that for ESD is around 300 MHz.

4.1 Introduction

Shielding and filtering are two complementary practices. There is hardly any point applying good filtering and circuit design practices to guard against conducted coupling if there is no return path for the filtered currents to take. The shield provides such a return, and also guards against direct field coupling with internal circuits and conductors. Shielding involves placing a conductive surface around the critical parts of the circuit so that the electromagnetic field that couples to it is attenuated by a combination of reflection and absorption. Enclosures, cables, circuits and PCBs are the parts that are to be shielded. The shield can be an all-metal enclosure; if protection will be enough then a thin conductive coating deposited on plastic is sufficient.

The question arises – Why do we shield? The two reasons for shielding are as follows:

• To protect the victims (such as sensitive circuits) from radiated emissions. In many cases victims can be humans (biological hazards). There is an on-going debate on the damage to humans by RF fields. Figure 4.1 shows the chart that gives some guidance on exposure levels for different frequencies.
• To reduce interference from culprits (such as noisy circuits, transmitters).

Shielding is often an expensive and difficult-to-implement design decision, because many other factors (such as tooling, aesthetic, accessibility) work against it. Shielding on a PCB/circuit level does not cost much but shielding on a larger level is usually expensive compared to changing a PCB or even adding a few filters in the design stage. The figures mentioned below shows how shielding of large enclosures adds exorbitant costs. It shows the part that is shielded along with the shielding cost in US dollars.

• Individual IC – US$ 0.3
• Segment of PCB – US$ 1.00
• Whole PCB – US$ 10.00
• Sub-assembly – US$ 20.00
• Whole product – US$ 100.00
• System – US$ 1000.00
• Room – US$ 10000.00
• Building – US$ 100000.00

A decision on whether or not to shield should be taken as early as possible in the project. Interference coupling is via interface cables and direct induction to/from the PCB. You should be able to calculate to a rough order of magnitude the fields generated by the PCB tracks and compare these to the desired emission limit. If the limit is exceeded at this point and the PCB layout cannot be improved, then shielding is essential. Shielding itself does not affect common mode cable coupling and so if this is expected to be the dominant coupling path a full shield may not be necessary. It does establish a clean reference for decoupling common mode currents too, but it is also possible to do this with a large area ground plate if the layout is planned carefully.

A description of shielding issues can be best classified into two parts:

• The theory of electromagnetic (EM) attenuation through a conducting barrier of infinite range, and
• The degradation of theoretically achievable shielding effectiveness by practical forms of shield construction

4.2 Shielding and Murphy’s Law

From the above section, it is clear that shielding costs much more with size. It is important to make the product cost effective, which suggests – avoid shielding if possible. If the layout allows concentrated interface, a ground plate may be adequate with partial shielding. Design for the possibility of shielding being required with a variety of shields in mind i.e., with many options of shielding in hand.

Murphy’s Law mentions that the shields should have thickness greater than 0.5 mm and should be rectangular in shape. Shields should be as large as possible (the design permitting), and use different, irrational side lengths.

4.2.1 Skin effect and skin depth

The important characteristic of the barrier material is its skin depth δ. This is the distance into the material at which the current density has reduced to 1/e (0.37 or 8.7 dB) due to skin effect. For every distance δ into the material the current density drops by 8.7 dB (approx. 9 dB). The thicker the wall, the greater is the attenuation of the current through it. This absorption loss depends on the number of skin depths through the wall. The skin depth is an expression of the electromagnetic property that tends to confine AC current flow to the surface of a conductor, becoming less as frequency, conductivity or permeability increases.

Skin depth (δ) = (π.F.μ.σ)^-0.5 meters

where, F = frequency,

σ = conductivity,

μ = permeability

Figure 4.2 shows the skin depths of copper, aluminium and steel. For example, skin depth in aluminium at 30 MHz is 0.015 mm. This explains why thin conductive coatings are effective at high frequencies – the current flows only on the surface, and the bulk of the material does not affect the shielding properties. It is therefore possible to reduce the current density by 18 dB in material with twice the thickness (i.e., with two skin depths), and 27 dB if it’s thrice the thickness (i.e., with three skin depths), and so on. The requirements for an effective shield are high conductivity for electric fields and high permeability for magnetic fields.

4.2.2 Shielding principles

Applying the concepts of shielding effectively requires an understanding of the source of the interference, the environment surrounding the source, and the distance between the source and point of observation (the receiver). If the circuit is operating close to the source (in the near, or induction-field), then the field characteristics are determined by the source. A circuit operates in a near-field if its distance from the source of the interference is less than the wavelength (λ) of the interference divided by 2π, or λ/2π. If the circuit is remotely located (in the far, or radiation-field), then the field characteristics are determined by the transmission medium. If the distance between the circuit and the source of the interference is larger than this quantity, then the circuit operates in the far field.

For instance, the interference caused by a 1-ns pulse edge has an upper bandwidth of approximately 350 MHz. The wavelength of a 350-MHz signal is approximately 857 mm (the speed of light is approximately 300mm/ns). Dividing the wavelength by 2π yields a distance of approximately 5 inches, the boundary between near- and far-field. If a circuit is within 127mm of a 350-MHz interference source, then the circuit operates in the near-field of the interference. If the distance is greater than 127mm, the circuit operates in the far-field of the interference.

Regardless of the type of interference, there is a characteristic impedance associated with it. The characteristic, or wave impedance of a field is determined by the ratio of its electric (or E-) field to its magnetic (or H-) field. In the far field, the ratio of the electric field to the magnetic field is the characteristic (wave impedance) of free space, given by Zo = 377 Ω. In the near field, the wave-impedance is determined by the nature of the interference and its distance from the source. If the interference source is high-current and low-voltage (for example, a loop antenna or a power-line transformer), the field is predominately magnetic and exhibits a wave impedance which is less than 377 Ω. If the source is low-current and high-voltage (for example, a rod antenna or a high-speed digital switching circuit), then the field is predominately electric and exhibits a wave impedance which is greater than 377 Ω.

Conductive enclosures can be used to shield sensitive circuits from the effects of these external fields. These materials present an impedance mismatch to the incident interference, because the impedance of the shield is lower than the wave impedance of the incident field. The effectiveness of the conductive shield depends on two things: First is the loss due to the reflection of the incident wave off the shielding material. Second is the loss due to the absorption of the transmitted wave within the shielding material. The amount of reflection loss depends upon the type of interference and its wave impedance. The amount of absorption loss, however, is independent of the type of interference. It is the same for near- and far-field radiation, as well as for electric or magnetic fields.

Reflection loss for plane waves in the far field decreases with increasing frequency because the shield impedance, Zs, increases with frequency. Absorption loss, on the other hand, increases with frequency because skin depth decreases. For electric fields and plane waves, the primary shielding mechanism is reflection loss, and at high frequencies, the mechanism is absorption loss.

Thus for high-frequency interference signals, lightweight, easily worked high conductivity materials such as copper or aluminum can provide adequate shielding. At low frequencies however, both reflection and absorption loss to magnetic fields is low. It is thus very difficult to shield circuits from low-frequency magnetic fields. In these applications, high-permeability materials that exhibit low-reluctance provide the best protection. These low-reluctance materials provide a magnetic shunt path that diverts the magnetic field away from the protected circuit. Examples of high-permeability materials are steel and mu-metal.

To summarize the characteristics of metallic materials commonly used for shielded purposes: Use high conductivity metals for HF interference, and high permeability metals for LF interference.

The frequency of the interfering signal is a critical concern when selecting EMI shielding devices. Low frequency magnetic waves in the 1 to 30 KHz range, for example, are most effectively shielded by absorbing the signals in permeable material. High frequency signals (30 KHz and above) are most effectively shielded by reducing entry windows and by insuring adequate surface conductivity to ground.

A properly shielded enclosure is very effective at preventing external interference from disrupting its contents as well as confining any internally-generated interference. However, in the real world, openings in the shield are often required to accommodate adjustment knobs, switches, connectors, or to provide ventilation. Unfortunately, these openings may compromise shielding effectiveness by providing paths for high-frequency interference to enter the instrument. The longest dimension (not the total area) of an opening is used to evaluate the ability of external fields to enter the enclosure, because the openings behave as slot antennas. Equation below can be used to calculate the shielding effectiveness, or the susceptibility to EMI leakage or penetration, of an opening in an enclosure:

where λ = wavelength of the interference and L = maximum dimension of the opening Maximum radiation of EMI through an opening occurs when the longest dimension of the opening is equal to one half-wavelength of the interference frequency (0-dB shielding effectiveness). A rule-of-thumb is to keep the longest dimension less than 1/20 wavelength of the interference signal, as this provides 20-dB shielding effectiveness.

Furthermore, a few small openings on each side of an enclosure is preferred over many openings on one side. This is because the openings on different sides radiate energy in different directions, and as a result, shielding effectiveness is not compromised. If openings and seams cannot be avoided, then conductive gaskets, screens, and paints alone or in combination should be used judiciously to limit the longest dimension of any opening to less than 1/20 wavelength. Any cables, wires, connectors, indicators, or control shafts penetrating the enclosure should have circumferential metallic shields physically bonded to the enclosure at the point of entry. In those applications where unshielded cables/wires are used, then filters are recommended at the shield entry point.

LF magnetic shielding

Shielding against magnetic fields at low frequencies is to all intents and purposes impossible with purely conductive materials. This is because the reflection loss to an impinging magnetic field (R_H) depends on the mismatch of the field impedance to the barrier impedance. The low field impedance is well matched to the low barrier impedance and the field is transmitted through the barrier with only a few dB attenuation or absorption. Fortunately, the requirements of the EMC Directive generally don’t extend to magnetic shielding at low frequencies, with the possible exception of some types of apparatus that may be susceptible to power frequency fields.

A high-permeability material such as mu-metal or its derivatives can give LF magnetic shielding by concentrating the field within the bulk of the material, but this is a different mechanism to that discussed above, and it is normally only viable for sensitive individual components such as CRTs or transformers. For an infinitely long cylinder in a DC field the shielding factor is:

S_M = μ/2 × t/d

where, μ = material permeability,

t/d = ratio of material thickness to cylinder diameter

Practically, there is a fall off of shielding performance towards the ends, but for distances inside the cylinder greater than the diameter the shield can be regarded as infinitely long; alternatively a high-permeability end cap can be used. Both welded seams and high intensity fields have the effect of reducing the material permeability μ. High flux densities tend to saturate the material; if this is likely then increase the material thickness or use a double shield with a nested construction, with a higher saturation flux density material facing the impinging field. For prototyping, you can use foil magnetic shielding material that can easily be worked by hand. Production shields should be properly fabricated and this needs to be done by a specialist.

4.3 Apertures and shielding effectiveness

The graph shown in Figure 4.3 shows how shielding effectiveness (SE) deteriorates with frequency and aperture dimension. A 100 mm aperture will roughly let through 20 dB more than a 10 mm one. Apertures are needed for ventilation, control and interface access, and for viewing an indicator. Seams, i.e., discontinuities at the joints between individual conductive members, also act as apertures.

Much academic effort has been spent in trying to quantify the general effect of apertures and discontinuities in shielding. Apart from the special case of sphere, a simple analytical expression based on the fundamental field equations has been unobtainable, leaving us until recently with recourse either to numerical computer modeling of a particular situation, or to various empirical rules of thumb. The possible number of variables that compound the difficulties are as mentioned below:

• The shape, size, position and number of apertures
• The relative positions and distances between the field source, the shield barrier and the apertures in it, and the victim
• The geometry of the shielding enclosure
• The frequency and source impedance of the field

All of these have to be taken into account by the model, even when considering the possible variations in the properties of the shielding material itself.

If adequate computing resources and skilled modeling staff are available, some shielding problems can be successfully modeled by numerical methods and substantial academic work has gone into improving these. Results have yet to be made available in a useable form to wider industry applications. This leaves us with some general guidelines and a few simplistic equations. The most widely quoted of these equations is

SE = 20 log (λ/2l) dB (below resonance)

where, λ = wavelength,

l = maximum aperture dimension

The above equation suggests that the shielding effectiveness (SE) degrades proportionally to frequency and is inversely proportional to aperture size until the aperture size is a half wavelength, at which point shielding is zero. This highlights the general dependence on frequency and size of aperture; i.e., the larger the aperture, the greater is its effect and the higher the frequency, the less is the available shielding. But as a means of predicting actual shielding effectiveness the equation is quite inaccurate, since SE also depends on the dimensions of the enclosure and the point at which SE is measured.

The lambda by twenty (λ/20) is a good rule of thumb for establishing maximum aperture size. For aperture(s) with the longest dimension of λ/20, the shielding effectiveness (SE) will reduce by 20 dB. Considering this situation, the distance between fastening screws, apertures, holes, etc. should be less than λ/20.

4.4 Waveguides

If necessary, improved shielding of vents can be obtained at the expense of thickness and weight by using honeycomb panels, in which the honeycomb pattern functions as a waveguide below cut-off. Figure 4.4 shows the dimensions of a waveguide. Figure 4.5 shows the graph of Shielding Effectiveness (SE) vs Frequency. Waveguides have better shielding effectiveness (below the cut-off frequency) than two-dimensional apertures.

4.5 Gasketting and sealing

Closer spacing of fixing points gives better control of higher frequencies and/or higher powers, although if more frequent bonds are required it may be easier to slip a length of a suitable EMC gasket material between the mating surfaces. The main parameters with gaskets are:

• Mechanical compliance
• Corrosion resistance
• Operation rate/day before becoming ineffective
• Environmental sealing, etc.

The pressure of the fixings should be sufficient to squeeze the gasket over the entire distance between the fixings with a pressure in excess of its minimum specification. Where the metalwork is less than sturdy, the pressure of the gasket material may make it bow between the fixing and look untidy. Worse than this, the metal may be bowed so much that a gap is opened up and the purpose of the gasket defeated. Additional strengthening plates may be found necessary to prevent this bowing, or else a lower-pressure gasket material could be used.

A very wide variety of gasket materials and products are now available, some of them at low cost, and some with additional chemical or environmental properties. To achieve good contact without excessive pressure, Beryllium-copper or stainless steel spring fingers may be used instead of foam or mesh type gaskets. Care must be taken to ensure that the gasket materials are compatible with the metal types being bonded, taking account of the environment (condensation, spray, corrosive gases, etc.) to ensure that the electrical bonds last the life of the installation.

4.6 Panel displays and keyboards

As shown in Figure 4.7, a dirty box can be constructed to reduce radiation from panel displays. Note that the connections to the display are filtered when going through the clean/dirty box barrier. If a system does not have a clean/dirty box construction, it will typically require a mesh in front of a display to attenuate unwanted radiation. Eg – a Digital Panel Meter as shown in figure below.

4.6.1 Shielding displays

Figure 4.8 shows the arrangement for shielding displays. The different layers are as shown below, the wire mesh will let through light but attenuate RF. Note that an EMC gasket must be used to keep radiated noise from creeping around the mesh.

4.7 Ventilation and shielding

One application for waveguide tubes is to provide ready-made ventilation panels for shielded enclosures. Since the walls of the waveguide can be made quite thin and the tubes can be stacked together with little effect on the electromagnetic attenuation, an assembly of such tubes can be stacked together with little effect on the electromagnetic attenuation. An assembly of such tubes can be built to provide a high degree of through airflow as well as high level of attenuation. These are available as pre-packaged units known as honeycomb panels which can be simply be fitted into the wall of an enclosure to give any reasonable level of the same open area in a thin panel, but of course are more costly and require some thickness in addition to the panel. The wire mesh approach may be adequate for many low-performance applications. To avoid variations in attenuation due to wave polarization, it is common to stack two layers of honeycomb panel against each other, with different orientations of the waveguide tubes in each layer.

4.7.1 Shielding non-metallic enclosures

The conductive coating may be technically feasible. There can be problems at the joints; parts get distorted during tightening. Gasketting is also not very effective. Nickel is effective but there are problems with skin depths/coating thickness. At the same time shielding of electronics should be considered.

Figure 4.9 shows the shielding types/technologies. Conductive coatings can be an alternative to solid metal shields.

Cables entering and leaving an enclosure should have their screens properly terminated to the enclosure wall. This means that a full 360° contact should be maintained around the outer surface of the cable screen.

Mechanisms for ensuring this are similar to conventional cable glands for environmental sealing, except that the appropriate parts are fully conductive. Most of the traditional manufacturers of cable accessories are now aware of the importance of EMC aspects, and provide EMC-specific cable glands as part of their stock range. A typical construction using tapered washers to compress an iris-type spring against the cable screen outer surface. This is one of the most common methods of clamping to the screen, but others are possible, including collect or other clamping mechanisms, elastometric compression modules, or folding the screening braid back over a conductive tube.

Aspects that you need to consider when specifying a screening cable gland system are:

• Mechanical compatibility – the cable screen outer diameter must match the gland’s construction, often within quite tight tolerances
• Electrochemical compatibility – the materials used for screen, gland and enclosure panels should discourage corrosion
• Ease of assembly, especially if unskilled or poorly trained technicians are expected, or the working area is restricted
• Conductivity across the joints, which directly affects shielding effectiveness
• Least disturbances of the screen
• Whether or not an environmental seal is also required

4.8 PCB-level shielding

It uses principles similar to shielding modules.

• No holes in the shielding
• Use filters at the interface
• Consider the reference plane as a part of the shield
• Bond the shield to the reference plane at regular intervals
• Consider guarded striplines at entry/exit points (equivalent to screened cables)

Figure 4.11 is an example of shielding on a PCB. Note that there are holes in the shield, but it does not affect shielding effectiveness (SE) if they are small enough.

5.1 Introduction

There has always been confusion between the terminologies Earthing and Grounding. In conventional electrical usage, Grounding is American for the same function as Earthing is in English; i.e., a safety protective function. But the EMC functions are different; the word Grounding will be used to distinguish the EMC function while Earthing will be used for the safety function.

The recognized and accepted purpose for grounding is to give a reference for external connections to the system. The classical definition of a ground is an equipotential point or plane that serves as a reference for a circuit or a system. Unfortunately this definition does not carry any meaning in the presence of ground current flow. Even when signal currents are negligible, induced ground currents due to environmental magnetic or electric fields will cause shifts in ground potential. A good grounding system will reduce these potential differences by comparing the circuit operating levels, but it cannot completely discard them. It has been suggested that the term Ground (as conventionally used) should be replaced with reference point to make the purpose of the node clearer.

An alternative definition for a ground is a low impedance path by which current can return to its source. This emphasizes current flow and the consequent need for currents always circulating as part of a loop. The task here is to design the loop in such a way that induced voltages remain low enough at critical places. Designing the ground circuit to be as compact and as local as possible is the only way to do this. The primary EMC function of a ground system is to minimize interference voltages at critical points compared to the desired signal. For this, it must present a low transfer impedance path at these critical locations.

The major part of the problem has been that different practices are suitable for different purposes; even if the purpose is purely EMC, frequency range is an important parameter, and its relevance is generally misinterpreted.

5.2 Earth and safety ground

Grounds fall in to two main categories:

• Safety ground (also called Earth)
• Signal ground

Signal grounds may or may not be at Earth potential, but safety grounds have to be at Earth potential. One ground should not be used for both signal ground and safety ground.

Figure 5.1 shows how disconnecting earth wires may cause shock hazards. This can happen with a so-called clean ground. A Clean ground can be good from a noise point of view, but unsafe.

Before the IC chip revolution of the ’80s most systems were connected to a common single ground system. This had the advantage of being cheap, efficient and safe. Since all devices were (theoretically) at the same voltage level (0 volts), a person touching two devices would not get ‘shocked’. This system of connecting devices together and then to a single common ground point was abandoned in the late 1980s and 1990s due to excessive noise being transmitted through the common ground into sensitive microcontroller devices. Terms such as clean and digital ground came into use. These grounds were often completely separate from dirty ground systems. It was found in the 1990s that although this system of separate grounds was good from a noise position, it was considered unsafe. This is because by separating the grounds the voltage level of two or more devices would float. This could possibly make devices have difference voltages and therefore render them unsafe.

Figure 5.2 shows a safe setup with the enclosure grounded/ earthed. All device grounds are required to be connected together at one common point. To reduce the noise induced into sensitive IC equipment the quality of the ground must be increased over the simple single point grounding systems of the past. By putting in grid grounds and parallel common point ground the noise induced into devices from other commonly connected grounds can be reduced. Also, since the devices are connected together their ground voltage levels should be at the same level and therefore safe.

With the boost in the use of mobile transmitting devices, EMI safety has taken on a new dimension. Mobile telephones have been known to cause fires at gas stations, turn devices on and cause machinery to act unpredictably. This problem has the potential to cause catastrophic failures in systems, which in turn could cause major accidents, injuries and death.

5.2.1 Ground types

One ground does not essentially fit all. Different types of grounds have different current and frequency requirements. Figure 5.3 shows the types of grounds along with their corresponding Bandwidth, Time scale and Level.

Type	BW	Level	Timescale
Power safety	50/60 Hz	10 to 1000 A	Sec to min
Lightning	1 MHz	Upto 100 kA	msec
ESD	300 MHz	10 to 50 A	nsec
EMI	DC to daylight	μA to A	Nsec to years

5.3 Grounding and frequency

Different types of techniques are used for grounding. Some of them are used at low frequency while others are used at high frequency. These can be classified as follows:

• Low frequency phenomena/technique
• High frequency phenomena/technique

Figure 5.4 shows the range of frequencies by which these techniques are partitioned as low and high frequency techniques. The low frequency techniques ranges from DC (i.e., 0 Hz) to around 10 kHz, while high frequency techniques ranges from 10 MHz to GHz. The crossover region (i.e., between 10 kHz to 10 MHz) of what is considered as high or low frequency grounds can be related to the λ/20 rule. These techniques are a function of frequency, but sometimes they may sound conflicting, e.g., on one hand, single point grounding and on the other hand, planes or grids.

5.4 Ground loops

Modern day industries and businesses rely largely on electronic systems. These include industrial drives, distributed control systems, computer systems and networking equipment or communication electronics. These electronic devices often work with very low power and voltage levels for their control and communications and cannot tolerate even small over voltages or currents. The result of such voltage/power surges is invariably the failure of the electronic device itself. It is also possible that induced voltages from nearby power circuits experiencing harmonic current flow can cause interference in the systems carrying communication signals and can result in malfunctions due to erroneous signal transmission. Due to this sensitive nature of electronic and communication equipment, any facility that houses such equipment needs to have its electrical wiring and grounding systems planned with utmost care so that there are no unpredictable equipment failures or malfunction.

One of the important and main factors of concern is the noise due to Ground loops. Figure 5.5 shows the illustration of how a ground loop is formed between the grounds of two circuits (circuit 1 and circuit 2).

Ground loops are problems because currents return in the ground circuit by separate paths. Separate paths have different impedances and cause ground voltages that shift with respect to each other. This shifting of the grounds is seen as noise in the circuitry, and therefore a practice has been developed of trying to discard all such loops. This practice may although often be superficially successful but is unfortunately misguided.

Consider a situation where high earth potential differences exist, and closing a loop between two such earth points will allow a high current to flow in the structure. If the conductors in that loop include a segment that either forms part of, or is closely coupled to a signal or low-level power cable, then substantial interference will be induced in that cable. If the loop is not closed (i.e., opened), the current no longer flows, and hence no interference – although the high potential differences still remain and is ready to create interference as soon as another loop is formed somewhere else. This is the principle that is now a convention in the star or single point earth regime – remove all ground loops and live with the resultant high voltages between different parts of the earthing system.

In low frequency systems, such an approach is fairly easy to implement and at the same time is also quite successful, but it represents a retreat from best practice. Now that interference is typically measured in the MHz region rather than Hz, it is untenable. This is because earthing conductors provide high impedance at these frequencies and therefore they tend to de-couple a system from earth, rather than couple to it. Another effect of ground loops is that they act as magnetic/inductive loops (or antenna). Magnetic flux couples through the loop and induces a noise voltage that shifts the ground.

In these circumstances the only reliable and efficient earthing system is a mesh. The mesh provides a multiplicity of ground loops (but they are small and controlled) – potential differences between parts of the structure are reduced, resulting currents are low and the interference consequences (if any) are negligible. The safety, reliability, ease of use and EMC advantages of properly implemented MESH-BN systems by far outweighs their possible disadvantages.

5.5 Ground impedance

Earthing wires do not offer end-to-end low impedance at high frequencies. Even at moderate frequencies, their impedance is quite significant. Ground impedance depends on the frequency:

• At DC (and low frequency) the impedance is purely resistive.
• In the medium frequency range, the impedance is inductive. The impedance of the length of wire increases linearly with frequency as long as the inductance dominates.
• At higher frequencies, transmission line and antenna effects are evident.

A piece of wire is not a high frequency connection, use bonding straps instead as it works because of its large area. At high frequency, current crowds towards the surface of the conductor (skin effect), for this current to have low impedance, it must flow freely across the large area.

5.6 Ground topologies

Grounding systems for a circuit reference can be configured by either of the below mentioned three ways:

Single point ground – the single point grounding system is conceptually the simplest, and it reduces common impedance ground coupling and low frequency loops. Figure 5.6 shows the single-point ground configuration.

Each rack or sub-unit has one bond to the chassis, and each circuit module has its own connection to a single ground. Any currents flowing in the rest of the ground network do not couple into the circuit. Such a system works well up to frequencies in the MHz region, but the distances involved in each ground connection means that the common mode potentials between circuits start to build up as the frequency increases. The circuits are effectively isolated from each other at distances of odd multiples of a quarter wavelength and at the same time, stray capacitance starts to contribute to sneak current- paths that bypass the single point wiring.

Multi-point grounding – Figure 5.7 shows the multi-point ground configuration.

Multi-point grounding and hybrids can overcome the RF problems associated with the pure single-point systems. Multi-point grounding is mandatory for digital and large signal high-frequency systems. Circuits and modules are bonded together with many short links to reduce ground-impedance-induced common mode voltages. Alternatively, many short links to a chassis, ground plane or other low impedance conductive body are made. If the loops are formed then they may cause magnetic field pick-up, and hence this may not be suitable for sensitive audio circuits. It is difficult to keep the supply frequency (50/60 Hz) interference out of such circuits. But this is only a problem if the circuits themselves are prone to it and if amplitudes are large because the loop areas are large. Circuits that operate at high frequencies are not susceptible to this interference.

Hybrid grounding – In hybrid grounding systems, reactive components such as capacitors and/or inductors are used to make the grounding system act differently at low frequencies and at RF (high frequencies). With a hybrid ground a capacitor is used to complete the ground circuit. At low frequencies where ground loops are the problem, the capacitor will tend to isolate the grounds. At high frequencies, capacitors act as shorts and multi-point grounds result.

5.7 Guidelines for grounding

It is not practical to optimize the ground layout for all individual signal circuits, so circuits with the greatest threats should be concentrated on. These are the ones that most frequently carry the high di/dt signals (clock lines and data lines, and square wave oscillators at high powers, especially in switching power supplies). From a susceptibility point of view, sensitive circuits (such as edge-triggered inputs, clocked systems, and precision analog amplifiers) should be treated similarly. After identification and partition of these circuits, loop inductance and ground coupling are concentrated. Here are the guidelines made for grounding:

• Don’t compromise on safety grounds
• Analog circuit grounds
• Digital circuit grounds
• I/O grounds
• Ground Map

5.7.1 Grounding analog circuits

The prime objective here is to prevent noisy ground currents to interfere with sensitive analog ground paths. Analog circuits such as motors, relays etc. fall in the low frequency region and should be grounded by single star point. Extra measures, such as separate or filtered power and local grounds, should be taken depending on circuit sensitivity. Separate analog, digital and power grounds should be present and they can be joined at a single point generally at an A/D or D/A converter. If there are more than one A/D or D/A converters, use more than one ground plane (Figure 5.8).

5.7.3 I/O grounding

The prime objective here is to prevent the circuits from becoming unintended transmitters and receivers and to prevent the cables from becoming unintended antenna. Figure 5.9 shows that a digital ground plane should never be extended over an analog section, as this will couple digital noise in analog circuitry.

A single point connection between analog and digital grounds can be made at the systems A/D (Analog to Digital) converter. It is very important here to not connect the digital circuitry separately to an external ground, and if this is done then extra current paths are set up that permit digital circuit noise to circulate in the clean ground. Interfaces to the digital circuits (for instance, an input or output port) should be buffered so that there is no need to be referenced to the digital 0 V. The best interfaces available are an opto-isolator or relay, but these are expensive. When isolation cannot be afforded, a separate buffer IC that can be referenced to the I/O ground is preferable, otherwise buffer the port with series resistors or chokes and decouple the line at the board interface with a capacitor and/or a transient suppressor to the clean ground.

6.7.1 Cables into screened compartments

Anything that interrupts the current flow will affect the shielding effectiveness. Any discontinuity in the conductive material will interrupt the current flow. Discontinuities are due to a number of factors:

• Joints between structural members and/or panels
• Large openings for windows, access doors
• Small openings for ventilation holes, connectors, controls

The fastest way to destroy shielding effectiveness is to make a hole in it and run a cable through it. Figure 6.6 shows a good practice, but the best way to do it is by using a filter at the interface.

7.1 Introduction

EMC should be a part of each and every stage of the designing process. Although existing designs can be tinkered with, there is no alternative other than re-designing to ensure compatibility along with provisions for future changes (i.e., to make the necessary changes in the design whenever required). It simply means that there has to be a lot of effort made before the designing process is initiated (Think well before starting!!!). Such an action usually proves to be more effective and cheaper in the long run.

Now the question arises – How to start with a good design? The answer to this is to have knowledge and a better understanding of the hidden components involved in any schematic diagram. Take the example of Figure 7.1; this is the schematic that we can see.

Figure 7.2 shows the schematic involving hidden components (which we cannot see, but have to be considered for better design).

Now just see the difference between the schematics in Figure 7.1 and Figure 7.2. The former merely shows the output of an 8 input NAND gate connected to one of the inputs of a 2 input NOR gate (this is what we can see). The latter shows the other hidden components involved.

What are these hidden components?

The answer to this question is the stray capacitance and lead inductance. They can be explained as follows:

• Any conductor in the vicinity of any other conductor gives rise to parasitic (or stray) capacitance.
• Any conductor of finite length has some inductance along its length. This is usually called Lead Inductance.

Thus the presence of these hidden components in Figure 7.1 gives rise to Figure 7.2. These components not only change the look of the circuit but also change the overall operation (or behavior) of the circuit and so these components should never be ignored while designing a circuit. Utmost care should be taken while initiating a design process and all such hidden components should be included, otherwise they may hamper the design when put in practice. The circuit will not behave the way it is intended to, just because hidden components were not considered and one may not even understand the reason for such an unusual behavior of the circuit (even though it works fine on paper). And so it is said – thorough knowledge and understanding of these hidden components is mandatory for circuit designing. Even plain resistors, inductors and capacitors don’t behave as ideal components on a schematic.

7.2 Parasitic in passives

The parasitic elements present in passive components tend to change the role of the passive components and in turn they operate in a way that is not expected.

Some of them are mentioned below:

• Film resistors are capacitive (shunt capacitance of approximately 0.2 pF) and inductance (lead inductance and spiral tolerance); and because of the presence of this inductive and capacitive components, film resistors may resonate.
• On the other hand, wire wound resistors are useless above a few KHz.
• Capacitors have lead inductance as well as internal inductance, and so are mostly above resonance, they are inductive in behavior.
• Due to the presence of parasitic components there are limitations in power handling capability (i.e., the temperature co-efficient change), dI/dt (rate of change of current) capability and dV/dt (rate of change of voltage) capability that is generally found in tantalum objects.
• Stray magnetic fields (as in the case of transformers) and RF pick-ups are the effects that are observed in the case of magnetic components.

7.3 Clocking

The major source of radiation in digital circuits is the processor clock(s) and its harmonics. All the energy in these signals is accumulated at a few specific frequencies, with the result that the clock signal levels are 10 to 20 dB higher than the rest of the digital circuit radiation. Since the commercial radiated emissions standards do not distinguish between narrowband and broadband, these narrowband emissions should be reduced first – by proper layouts, grounding and buffering of clock lines. Then concentrate on the other broadband sources, especially data/address buses and backplanes, and video or high-speed data links.

It is usually seen that circuits are driven from clock signals whose transition times are substantially faster than required. Undoubtedly there are many high performance circuits where clock timings are critical and transitions must be as fast as possible, but this is not always the case. Wherever circuit constraints allow, slower clock rates tend to minimize harmonic generation. This can be done in three ways: series impedance, parallel capacitance or by using a low-performance buffer.

One possible alternative is a technique more often associated with radio transmission, known as Spread Spectrum clock generation. In this technique a waveform selected for the most even spectral-spreading modulates the clock frequency itself by 1.5 to 2 %. This results in a wider distribution of the spectral energy associated with each clock harmonic so that the level measured in a constant 120 kHz bandwidth falls, between 10 to 20 dB. This is easily achieved without any extra effort made in layouts and without slowing the clock rise times.

7.4 Choice of logic family

7.6.1 Isolation

Signals may be isolated at the input or output with either an opto-coupler or a transformer (Figure 7.5). The ultimate expression of the former is fiber optic data transmission, which with falling costs of components is becoming consistently more attractive in a wide range of applications. Knowing that the major interference coupling route is via connected cables, using optical fiber instead of wire completely removes this route. This leaves only direct coupling to the enclosure, and coupling via the power cable, each of which is easier to deal with electrical signal interfaces. Signal processing techniques will be needed to ensure accurate transmission of precision AC or DC signals, which increases the overall cost and broad area. Isolation is a good way of reducing common mode interference (especially at low frequencies). The advantage reduces as frequency increases (due to the barrier capacitance) until at very high frequencies, there is effectively no such thing as isolation.

Isolation quality is limited by the isolation of the power supplies. All isolation has some parasitic capacitances. At RF, it can be said that signal isolation is simply a myth.

It is generally found that common mode signals are the culprits. These can be removed with common mode chokes, baluns and isolation (Figure 7.6). If co-ax is used as shown in Figure 7.7, then it worsens emissions and increases susceptibility.

A specific problem with opto-couplers (that is not suffered by transformers) is that high-level but short transients can couple into the base of the receiving phototransistor and cause its saturation. This actually blinds the coupler, as long as it takes for the device to come out of saturation (which may be several μsecs); the overall impact on the circuit is to evidently stretch the duration of the transient to an extent that is not acceptable. The same effect happens with RF interference, and is due to capacitive coupling directly to the base (Figure 7.8), especially when this is brought out to a separate termination on the package. The problem is less with photodiode couplers, and can be made comparatively less severe with a phototransistor by connecting its base to an emitter with a low-value resistor (definitely at the expense of sensitivity).

7.6.2 Communication

Figure 7.9 shows the ways in which communications can be made. The first one shows co-ax, which is not good to use. Balanced transmission lines are far better to use (in comparison to co-ax). If they are screened, connect screen to earth rather connecting it to 0 V.

8.1 Introduction

In EMC work, filtering almost always means low-pass filtering. The purpose is to attenuate high frequency components while passing low frequency ones. The purpose of a filter in the context of EMC is to prevent interference from propagating either into or out of equipment via its interfaces. This reduces conducted coupling directly and also helps to reduce radiated coupling if the interference is radiated to or from the cables that connect to the interfaces. Filtering and/or transient suppression, across the appropriate frequency range for every unscreened cable that crosses the boundary, is necessary. This is simple to do for power cables, but for signal and control lines it may pose a problem. For this reason, it is often helpful to partition the whole system so that difficult to treat cables crossing the zone boundaries are kept at a minimum. Screened cables do not need filtering, although surge protection may occasionally be required on internal conductors. However, it is essential to bond the screen to the earth structure at the boundary, whether it is a screen wall or an earth bar. This ensures that the interference currents on the screen are returned to Earth rather than penetrating from one zone to another that would compromise the attenuation otherwise offered at the zone boundary.

Mains filter components are of course rated to withstand the nominal operating voltage plus some safety factor. But the mains supply frequently carries transients that cause an abrupt overvoltage of several times the nominal. Before selecting a filter, the considerations that are to be made are as follows:

• The likely transient environment and
• Whether the anticipated overvoltages will cause the filter too much stress.

When two non-conductive materials are rubbed together or separated, electrons from one material are transferred to the other. This results in the accumulation of triboelectric charge on the surface of the material. The amount of charge caused by this movement is a function of the separation of the materials in the triboelectric series (positive materials give up electrons more readily, negative materials acquire them more readily). Additional factors are the closeness, area of contact, and rate of separation.

The voltage to which an object can be charged depends on its capacitance, and follows the law Q=CV. The human body can be charged by triboelectic induction to several kV. Because a perfect insulator does not allow the movement of electrons, surface charges on an insulator remain in the area within which they are generated, but the human body is conductive and so a triboelectrically induced charge distributes itself over the body. The rate at which the charge will bleed off a body to its surroundings depends on the surface resistivity of the body and its surroundings. This in turn is a function of relative humidity; more the moisture in the air, lower the surface resistivity of insulators and hence quicker the charges bleed away. In practice, since movement is constantly generating charge, there is a balance between generation dissipation that results in a typical level of charge voltage that can be found in a particular environment.

When the body (in the worst case, holding a metal object such as a key) approaches a conductive object, the charge is transferred to that object normally via a spark. This happens when the potential gradient across the narrowing air gap is high enough to cause the spark. It is not mandatory that the target object be grounded. Charge transfer can occur between any two capacitive objects as long as there is a static potential difference between them, and a disruptive discharge current pulse flows.

8.2 Filter types and operation

From the above configuration it is clear that filtering is essentially the combination of providing a high series and low shunt impedance to unwanted signals. Undocumented characteristics of a variety of components at high frequencies and unanticipated source and destination characteristics can destroy a filter design.

R, L and C are the basic elements in the filter design. Figure 8.2 shows the various simple low-pass configurations and filter circuits are normally made up from a combination of these. Filtering can be as simple as inserting a resistor if the source and destination characteristics permit. The conventional low pass filter is built from two elements: series resistance or inductance, and a parallel capacitor.

The choice of resistance or inductance for the series element is usually determined by the signal current that has to be passed – power filters will typically be unable to stand more than a few ohms resistance, but signal filters might easily be able to cope with kilohms. Resistance has an advantage because it absorbs interference energy, does not contribute to resonance, and is of course cheaper and smaller than inductance. Inductance on the other hand can provide high RF impedance with little low frequency or DC loss.

The frequency range that has to be attenuated determines the choice of component types for a filter. Non-ideal components have parasitic impedance that limits their effectiveness. For example, capacitors have unwanted stray inductance and inductors have stray capacitance. High frequency performance demands that these parasitics are minimized, and this puts restrictions on the type of construction. In general, any single component can only achieve a maximum of 40–50 dB of attenuation before its parasitic elements limit its performance, hence high performance filters must use multiple stages. Capacitors create a low impedance to attenuate the interference frequencies and are therefore most effective in a high impedance circuit. Inductors create high impedance to the interference and are most effective in a low impedance circuit.

RC filters are the simplest and the most predictable filter. PI and TEE filters provide low loss to wanted signals and high attenuation to unwanted signals, but are resonant devices and depend on source and destination characteristics for their effectiveness.

8.3 Soft ferrites

One of the most common techniques for reducing both incoming and outgoing RF interference is the application of ferrite sleeves to cables at interfaces. The attractiveness of the ferrite choke is that it involves no circuit redesign, and often no mechanical redesign either.

Current flowing through a conductor creates a magnetic field around it. Transfer of energy between the current and the magnetic field is affected through the inductance of the conductor, for a straight wire the self-inductance is typically 20 nH/inch. In a non-permeable material (such as air or plastic sheath), the magnetic field strength and the magnetic flux density are equal. Placing a magnetically permeable material around the conductor increases the flux density for a given field strength and therefore increases inductance. Ferrite is such a material; its permeability is controlled by the exact composition of the different oxides that make it up (ferric, with typically nickel and zinc) and is heavily dependant on frequency. Also the permeability is complex and has both real and imaginary parts, which translate into both resistive and inductive components of the impedance inserted into the line passed through the ferrite. The ratio of these components varies with the frequency – at the higher frequencies the resistive part dominates and the assembly becomes lossy, so that RF energy is dissipated in the bulk of the material and resonances with stray capacitances are avoided or damped. It is very important to check ferrite impedance vs frequency characteristics for particular applications.

Figure 8.3 shows the effect of ferrites on differential mode (DM) and common mode (CM) signals.

The magnetic field produced by the intended go current in each circuit pair is substantially cancelled by the field produced by its equal and opposite return current, provided the two conductors are adjacent. Therefore any magnetic material (such as a ferrite sleeve) placed around the whole cable will be invisible to these differential mode currents. Placing a ferrite around a cable has no effect on the differential mode signals carried within it. A cable will also carry currents in common mode, i.e., all conductors have current flowing in the same direction.

When a ferrite is slipped over a single wire, it attenuates differential mode signals. On the other hand, when slipped over a pair of wires, it attenuates common mode signals.

8.4 Thumb rules

Capacitors work best for shunting noise from high impedance sources, while inductors work best for blocking noise from low impedance sources. From the various filter types mentioned in the above section, it is clear that the capacitor is always connected in shunt (whenever used) and the inductor is always connected in series (whenever used). Here are the thumb rules for various available conditions for unwanted signals:

• High impedance and DM – use shunt capacitor.
• Low impedance and DM – series inductor (both the legs at HF).
• High impedance and CM – identical shunt capacitors with each leg to local earth (usually chassis).
• Low impedance and CM – common mode inductors, all signal legs.

8.5 Filter specification

Before choosing a particular filter design, it is necessary to know the parameters that influence its design. Incorrect filter construction or mounting technique can easily compromise radiated emissions, and immunity. Poor shielding can easily compromise conducted emissions and immunity.

Filters should be located at the zone boundary. In addition to this, filters should be located as near as possible to the apparatus that is expected to be the source or victim of disturbances, to minimize the impedance of the connection. If as a compromise, the filter is positioned outside the protected area, the wiring between the filter and the protected area should be twisted and positioned close to the earthing structure.

The higher the frequency, the more a filter is compromised by RF leakage from its unfiltered side to its filtered side. For many engineers, the ease with which the RF leaks around a filter is surprising. Where an external cable to be filtered enters a shielded enclosure, the filter should be fixed into the metal wall at the point of cable entry and RF bonded to the metalwork all around the aperture it fits into. Through-bulkhead-filters are the best, since they maintain the integrity of the shield, but are often expensive to purchase and install. For mains supply filters, the IEC-320 inlet is the most common commercial style of bulkhead filter for up to 10 Amps, single phase, 230 Vrms.

Most mains filter manufacturers only specify their parts (because of commercial pressure) over the frequency range of the conducted emissions tests up to 30 MHz. The filter becomes less effective above 30 MHz and can compromise the shielding integrity of enclosures. This could possibly cause problems with radiated electromagnetic disturbances. Good layout inside the enclosure will minimize the high frequency coupling onto the internal filtered supply.

A filter must be designed to deal with the mode (DM or CM) that is most relevant to the interference; conversely, a filter that is attenuating the wrong mode will be ineffective. For example, a differential mode (DM) filter configuration will not affect common mode (CM) at all and vice-versa.

The filter designs should be flexible such that last minute changes can be implemented for more effective operation and at the same time, the cost of the design should be under control. All the above-mentioned points should be considered for optimizing the filter opertion.

8.6 Impedance matching

The performance of any filter depends on the impedance seen at its terminal. There are four relevant impedances for a simple single-phase mains filter:

• DM at the mains port
• CM at the mains port
• DM at the equipment port
• CM at the equipment port

All these impedances will be complex and frequency-dependant in real life. Conventionally, filters are specified for terminating impedances of 50 Ω at each end (source-end and load-end), because this is convenient for measurement and is an accepted RF standard. This leads us to a very important point that the filter specifications are optimistic when compared with their performance in reality.

Consider a typical supply filter, installed between the AC power supply and an AC-DC converter of the DC power supply of an electronics apparatus. The impedance of the AC supply can vary by as much as from 2 Ω to 2 kΩ depending on the loads that are connected to it, the nature of the supply transformer and the wiring to the point of connection. The impendence is complex and time- and frequency-dependent. Looking from the filter into the equipment, the impendence of the AC-DC converter circuitry appears as low impendence when the rectifiers are turned on near the peaks of the supply waveform, and as high impendence at all other times. The situation is far from being the matched 50Ω/50Ω setup used to measure filter attenuation.

Filter specifications employ a 50 Ω source and load impendence because most RF test equipment uses 50 Ω sources, loads, cables, and because the main specification standard (CISPR 17) requires this. For most practical uses of filters the specifications obtained by this method are at best optimistic, and at worst misleading. Filters made from inductors and capacitors are resonant circuits, and their performance and resonance can depend critically on their source and load impedances. An expensive filter with excellent 50/50Ω performance may actually give worse results in practice than a cheaper one with mediocre 50/50Ω specification.

Supply filters with a single stage are most sensitive to source load impedances. They can easily provide gain, rather than attenuation, when operated with source and load impedances other than 50 Ω. This gain usually appears in the 150 kHz to 1 MHz region and can be as much as 10 or 20 dB, leading to the possibility that fitting an unsuitable mains filter can increase emissions and/or worsen susceptibility. On the other hand, filters with two or more stages are able to maintain an internal circuit node at an impedance that does not depend very much on the source and load impedances, so they are able to perform better (at least nearly in line with their 50/50Ω specification). Of course, they are larger and cost more.

The easiest way to deal with the source/load impedance problem is to only use filters whose manufacturers specify differential mode (symmetrical) performance for both matched 50/50Ω and mismatched sources and loads. CISPR 17 requires mismatched figures that are taken with 0.1 Ω source and 100 Ω load, and vice versa. Now draw attenuation versus frequency curve consisting of worst-case figures from each of these various curves and use this as the filter’s specification. When filters are chosen using this technique (to try and meet the predicted or actual requirements) their performance can be as good as expected. When the 50/50Ω figures alone are used to predict filter performance the result is often disappointing. The performance will also be different depending on the dominant interference path, CM or DM, since the filter attenuation is different for each mode.

8.7 Filtering precision and Filter earthing

Differential mode interference is present between the conductors of the pair, and therefore the capacitor must be placed in parallel across the conductors and the inductor in series with just one conductor. On the other hand, the common mode interference appears on both conductors together, returning via the earth connection external to the desired circuit. Balanced transmission provides better immunity and so CM filters can be used.

All commercial filters are housed in metal bodies of some sort, with the body forming the filter’s earth connection. An IEC inlet filter with a metal body installed in a shielded enclosure can only give a good account of itself at frequencies above a few tens of MHz if its body has a seamless construction and its body is RF bonded to the shielding metalwork. The same is true for any other metal-bodied filter. The reason is that any inductance due to a wired earth connection will turn the filter into a high pass configuration. The greater the inductance (i.e., the longer the wire), the lower the frequency at which this becomes significant. Bonding the case directly to the chassis earth is the only sure way to realize a filter’s performance.

Adding further stages to the simple single-stage filter is usually the most cost effective way of improving a filter’s performance. Multi-stage filters are inherently less affected by the extremes of source or load impedance. Especially at AF, two-stage filters may be needed.

The filter is always located at a zone boundary. This is for two reasons:

? The filter is a part of the protection offered by the zone barrier. Locating it at a distance from the barrier would allow cables between the filter and the barrier to break this protection. ? A filter needs a high-integrity earth reference for good high-frequency operations. The zone barrier, which will usually be either a shielding wall in a cabinet or chamber or the entry to an earthing mesh, provides this directly. A large metal plate bonded to the earth structure at the SPC (Single Point of Connection) to a zone can also serve as such a reference.The safety earth (yellow and green wire) is not a reference point at a high frequency (HF) connection. The case’s potential with respect to the reference ground is defined by a complex network of inductances (connected cables) and stray capacitances, which are impossible to predict. As per the thumb rule, the inductance distribution along the track is 1nH/mm; the earthing efficiency determines the filter efficiency. Filters that are bonded to the ground plane are called Shielded; on the other hand, filters that are bonded to earth are called Unshielded.

Figure 8.4 shows the mechanics of filtration. Earthing effectiveness can be nullified by impedance to ground. A proper bonding strap should be used while grounding a filter for high frequency EMI.

8.8 Filters and shielding

A wire bridging a clean/dirty area can pick up noise from the dirty area and radiate it in a clean area. So cables crossing a partitioning zone need to be filtered. Any cable through a shield offers a potential leak point for EMI. Using a filter in a dirty box shield offers an excellent way of maintaining the shielding integrity combined with the use of filters. Figure 8.5 shows how to mount a screw-terminal filter using the dirty box method.

This encloses the filter in an individually shielded, segregated box within the main shielded enclosure. In clean/dirty box arrangement, the dirty box segregation is applied to just one filter. The filter input and output cables in the dirty box must be very short and far away from each other, but even so high frequencies may still leak across and ferrite sleeves may be needed on either or both cables if performance above 30 MHz appears to be compromised.

8.9 Surge protection devices (SPDs)

Surge arrestors are variable resistance devices, whose resistance is a function of applied voltage. They are designed so that they provide a clamping effect when the voltage across them exceeds a certain level.

There are four basic types of SPD:

• Gas discharge tube (GDT) – essentially just a spark gap, slow but very high power and negligible leakage.
• Metal-oxide varistor (MOV) – a bulk semiconductor, fast, less rugged, cheap and low clamping voltage than a GDT.
• Avalanche diodes – a semiconductor device with a zener type of action, fastest but not very high powered, lowest clamping voltages and most expensive to buy.
• Thyristor devices – another type of semiconductor device, slow but will handle high currents.

The GDT and thyristor devices have to be self-triggered before clamping, and during the initial period they let through surge voltages that may be potentially damaging. They also have a foldback characteristics, which means that once they fire-off and are carrying current, the voltage across them drops to well below the level they were previously able to block. Careful design is needed to make sure that, if connected to a source of DC current, they do not remain in continuous conduction, once triggered by the surge. This characteristic is not shared by the MOV and avalanche devices, but their disadvantage is a higher clamping voltage, and hence dissipation, for a given normal system operating voltage.

Figure 8.6 indicates the voltage V/s time waveform of the different types of SPD when exposed to the leading edge of a typical surge test waveform.

8.10 Surge protection and data integrity

In properly earth-meshed and light-protected structures, the central volume is least exposed to the effects of lightning. So this is the best place to install the most sensitive apparatus. Installation of electronics should be avoided on roofs, on top floors of buildings (especially tall), near to outside walls, near to outside corners, near to down conductors or near to tall structures (like chimneys). However in all metal structures resembling screened rooms the location of equipment is not generally of any concern. SPD techniques are needed because of the long cable run of telephone and data lines outside building structures and poor earthing techniques.

The general belief is that the interference caused in analog data can simply change the waveshape, which can be filtered and the signal is retrieved without much difficulty. But this is not always true; it will also depend on the frequency of the signal. As far as the software part goes, data integrity should not be taken for granted. It is the role of the software programmer to take into account the data integrity on the line, if the data is send or received over the line. Cyclic redundancy check (CRC) is one of the simplest protocols used.

Typical surge and ring waveshape for testing against the indirect effects of lightning are shown in Figure 8.7 and Figure 8.8 respectively.

8.11 Surge protection ratings

Commercial surge protection identifies three distinct zones with differing surge exposure categories: C (most reserve), B, and A (least severe).

• Category C – power conductor outside a structure; supply side of main incoming LV distribution board/switchgear; load side of distribution board/switchgear for outgoing mains cables such as to another structure or external apparatus such as transformers, pumps, external lights, etc.
• Category B – power conductors inside a structure; between load side of incoming distribution board and supply side of socket outlet or fused spur, or within apparatus not fed from a wall socket, or sub distribution boards located within 20 meters cable run of Cat C, or plug-in equipment or fused spur within 20m cable run of Cat C.
• Category A – power conductors to plug-in equipment or fused spur located more than 20 meters cable run from Cat C and/or 10 meters from Cat B (category A may not exist in smaller structures).

These values vary hugely from standard to standard. The IEC 1024 set of standards calls Category C, Zone ‘0’; Category B, Zone ‘1’; and Category A, Zone ‘2’. Figure 8.9 shows the various zones.

All data/signal cables (e.g. telephone wires) entering a structure from outside are Category C over their whole length, except internal to the structure after suppression by SPDs, since surges in these lines are time-stretched over distance, rather than attenuated as happens with power cables. BS6651 substantially agrees, but suggests an upper limit of 20 kA in Category C in high-risk areas. The exposure levels suggested by BS6651 are based on lightning risk assessment only. If transients of other origin are present, consider upgrading any SPDs and/or specifying apparatus with higher surge immunity. For instance, if in an industrial area the risk assessment suggests that an SPD suitable for medium exposure level is appropriate, the presence of inductive switching transients may make a high exposure level SPD more appropriate. In these circumstances, specialist advice should be sought.

8.12 Surge protection fusing

All SPDs fail eventually, and since the majority use metal-oxide-varistor types (whose failure mode is sustained leaking and finally short-circuiting), those that are placed across power conductors need to be fused. The choice to be made is whether protection of the load (equipment), or continuance of operation is the most important parameter, and is a user’s decision.

If the fuse is in common line that also goes to the load (protected equipment) (see ‘a’ in the Figure 8.10), then the opening of the fuse due to SPD failure will remove the power from the equipment, which may not be acceptable in some applications.

If the fuse is dedicated to the SPDs alone (see ‘b’ in Figure 8.10), when it opens during a surge event that kills the SPD, the load (protected equipment) may be exposed to the remaining parts of the surge and damaged. Afterwards, even if the load (protected equipment) is undamaged, it has lost its surge protection and so is very exposed to the next surge that comes along.

The arrangement of Figure 8.10 (b) is preferred for domestic premises, where no inspection and preventative maintenance of SPDs can be expected, since the continue opening of fuses will inform the user that there is something wrong that he should get fixed. Where a system of inspections and preventative maintenance is in place, and if recent models of SPDs with health indicators are fitted, either fusing arrangement may be chosen.

8.13 Positioning SPDs

SPDs are connected directly across the line to be protected. They are first installed at the equipotential bonding plate, or main earthing terminal in less sensitive structure on all incoming or outgoing conductors that are not directly bonded to the earth at the same place. Other SPDs may be fitted at lightning protection zone boundaries in a similar way within the structure, and some may be fitted at the equipment itself. In all these arrangements, the common need is to keep their lead lengths short and to ensure that their earthing is relative to what is being protected. Connecting to a separate SPD earth is less productive and nullifies protection.

The total let-through voltage of an SPD is the sum of its own voltage clamping action and the transient voltage drop in its connecting leads due to their inductance (usually 1 nH/mm, rule of thumb). For the rate of change of current associated with lightning strikes, if an SPD has an earth lead longer than 1000 mm, greater voltage surges can be let-through to the equipment being protected. For example, in a 6 kV–3 kA test an SPD gave a let-through of 630 volts when it was installed correctly (where a lead’s length is less than 250 mm and bound together), but 2000 mm long bound leads, let-through a total 1,200 V and unbound 2000 mm leads let-through 2,300V. In general, SPDs for lightening suppression should have a lead length of 500 mm or less. Although large diameter wires are slightly better than thinner ones, the most important concern is to achieve the shortest lead length. Binding the leads to/from an SPD also helps reduce lead inductance and in turn let-through voltage. Always make sure that SPD protected cables are segregated from any cables that have not been protected SPDs by at least 150 mm on parallel runs.

When SPDs are correctly fitted to signal and data lines of equipment, their earth bonds will have short connections to the chassis (local earth) of the equipment itself. But the neutral conductor in mains supply is earthed many meters away (for other good EMC reasons), so if the surge is clamped by the signal’s SPD, it will inject a current into the chassis and hence there is a considerable hike in voltage with respect to the remote neutral earthing point. This could create a surge voltage problem for mains supply. A similar problem can occur when SPDs are fitted locally to the equipment’s mains terminals, with the resulting over voltage appearing on its communication cables. So when fitting SPDs to any cables of equipment, it generally means that all inputs, outputs and power cables need to be fitted with SPDs too. Evidently this could be very expensive for equipment with a large numbers of cables.

8.14 Electrostatic discharge

Electrostatic discharge is a source of transient upset that typically occurs when a person or other body that has been charged to a high potential by movement across an insulating surface then touches an earthed piece of equipment, thereby discharging through the equipment. Currents of tens of amps can flow for a short period with a very fast risetime (in the order of few nanoseconds). Even though it may have low energy and be conducted to ground through the equipment case, such a current pulse couples very easily into the internal circuitry and can disrupt its operation. When two objects with a high electrostatic potential difference approach, the air gap between them breaks down and the charge difference is equalized by the current flow across the resultant ionized path. The current route is completed by stray capacitance between the objects and their surroundings, and the inductance and resistance of the path.

The actual current waveform is complex and depends on many variables, which includes speed and angle approach, environmental conditions, as well as the effect of the distributed circuit reactances. At lower voltages a precursor spike due to the local area of the source (a finger or a metal tool) is produced which has a very fast risetime, of the order of a few hundred picoseconds. This spike, although low in energy, can be more damaging to the protection of fast digital equipment than the bulk discharge that follows it, which may have a risetime of 5–15 nanoseconds. The discharge current will take the route of least inductance. If the case is well bonded to ground then this will be the natural sink point. If it is not, or if it is non-conductive then other routes may be via the connecting cables. Because the discharge edge has an extremely fast risetime (sub-nanosecond), stray capacitive coupling is essentially transparent to it, whilst even short ground connectors of a few nH will present high impedance. Apertures in a conductive enclosure will also result in intense local magnetic fields which couple to internal circuits.

The discharge current pulse di/dt and its indirect effects produce the principal effects of an ESD in terms of equipment malfunction. The rate of change of electric field dE/dt when the local static charge voltage collapses can also couple capacitively into high-impedance circuits; in some circumstances into the high static electric field itself, before a discharge happens. This may cause undesirable effects.

The resultant sub-nanosecond transient equalizing current of several tens of amps follows a complex route to ground through the equipment and is very likely to upset digital circuit operations if it passes through the circuit tracks. The paths are defined more by stray capacitance, case bonding and track or wiring inductance than by the designer’s intended circuit. The high magnetic field associated with the current can induce transient voltages in nearby conductors that are not actually in the path of the current. Even if not discharged directly to the equipment, a nearby discharge such as to a metal desk or chair will generate an intense radiated field that will couple into unprotected equipment. Critical areas that can act as sink points for the ESD are exposed metalwork, apertures, front panel components and connectors. Components and apertures can allow a discharge to take place via creepage across a surface to the circuits inside an enclosure, even if the enclosure itself is insulated. The breakdown voltage gradient in dry air is approximately 30 kV/cm but his is reduced considerably across a surface, especially if the surface is contaminated with dirt or other substances.

For reduced ESD susceptibility, circuit ground needs to remain stable during the ESD event. A low inductance ground network is essential, but this must also be coupled (by capacitors or directly) to a master reference ground structure. I/O cables and internal wiring may provide low-impedance paths for the current in the same way as they are roots into, and out of the equipment for common mode RF interference. The best way to eliminate susceptibility of internal harnesses and cables is not to have any, through economical design of the board interconnections. External cables must have their shields well decoupled to the ground structure. Insulated enclosures make the control of ESD currents harder to achieve, and a well-designed and low inductance circuit ground is essential. But if the enclosure can be designed to have no apertures which provide air gap paths to the interior then no direct discharge will be able to occur, provided the material’s dielectric strength is high enough. You will still need to protect against the field of an indirect discharge, though.

Keyboards present an operator interface that is frequently exposed to ESD. Keyboard cables should be foil-and-braid shielded which are 360°grounded at both ends to the low inductance chassis metalwork. Plastic key caps will call for internal metal or foil shielding between the keys and the base PCB that is connected directly to the cable shield, to divert transients away from the circuitry. The shield ground should be coupled to the circuit ground at the cable entry point via a 10–100 nF capacitor to prevent ground potential separation during an ESD event. A membrane keypad with polyester surface material with an inherently high dielectric strength and is therefore resistant to ESD but it should incorporate a ground plane to provide a bleed path for the accumulated charge and to improve RF immunity. The best way to avoid the effects of ESD is to keep plastic enclosures, membrane keyboards, plastic knobs, plastic pot shafts, plastic switch caps, plastic lenses, etc away from each other as much as possible.

8.15 Creepage and clearance

The mechanical design of many electrical products relies on separation distances between live and accessible parts to maintain safety against shock. When such products are placed within screened enclosure, it is necessary to ensure that such separation distances are not compromised. A typical problem might occur when a product in a plastic case with normally sufficient separation between hazardous and safe circuit connections, is bolted to a metal frame inside a screened enclosure; the separation distances from each set of connections to the metal frame may not now be sufficient. A related issue occurs when a plastic case is conductively coated for screening purposes. The original designer may well have relied on the internal case geometry being insulating to maintain safety clearances. Adding a coating can breach these clearances by providing a new conducting path. The dangers of coating also extend to the possibility of flakes of coating becoming detached due to environmental stresses such as vibration or temperature extremes, and lodging in areas where they may bridge critical insulation. Screening is not the only situation in which clearances may be affected: a poorly designed or installed mains filter or surge suppressor can create similar issue.

Dielectric breakdown of air is around 30 kV/cm. With the human body having the ability to acquire a charge of 20 kV, any creepage distance shorter than say 7 mm can lead to a discharge reaching circuitry.

8.16 Shielding

The safety bond, a length of green/yellow wire interconnecting panel and frame or different panels, is familiar to system designers. But it is important to realize that this is not sufficient for EMC bonding. The purpose of a wired safety bond is to prevent different parts of the structure from assuming different potentials and hence presenting an electric shock hazard at power frequencies (it has no other purpose). It cannot give a low impedance connection at RF.

Signals may be isolated at input or output with either an opto-coupler or a transformer. The best example of the former is fibre optic data transmission, which with falling costs of components is becoming steadily more attractive in a wide range of applications. Isolation breaks the electrical ground connection and therefore substantially removes common mode noise injection, as well as allowing a DC or low frequency AC potential difference to exist. However there is still a residual coupling capacitance between primary and secondary that will compromise the isolation at high frequencies or high rates of common mode dv/dt. This capacitance is typically 2-3pF per device for an opto-coupler; where several channels are isolated the overall coupling capacitance (from one ground to the other) rises to several tens of pF. This common mode impedance is a few tens of ohms at 100 MHz, which is not much of a barrier.

An electromagnetic shield is normally made from several panels joined together at the seams. Unfortunately, when you join two sheets the electrical conductivity across the joint is imperfect. This may be because of distortion, so that the surfaces do not fix well, or because of painting, anodizing or corrosion, so that an insulating layer is present on one or both the metal surfaces. As a result of this weak spot (shield joint), the shielding effectiveness is reduced. Shielding effectiveness is defined as the ratio of the field strength impinging on a shielding barrier to the field strength propagated away from the other side of the barrier. It can be expressed as the sum of reflection, absorption and re-reflection losses, and is given by the equation below:

SE (dB) = R (dB) + A (dB) +B (dB)

An electric field impinging on a conductive wall of infinite extent will induce a current flow in that surface of the wall, which in turn will generate a reflected wave of the opposite sense. This is necessary in order to satisfy the boundary conditions at the wall, where the electric field must approach zero. The reflected wave amplitude determines the reflection loss of the wall. Because shielding walls have finite conductivity, part of this current flow penetrates into the wall and a fraction of it will appear on the opposite side of the wall, where it will generate its own field. The thicker the wall, the greater the attenuation of the current through it. This absorption loss depends on the number of skin depths through the wall. The skin depth is an expression of the electromagnetic property that tends to confine AC current flow to the surface of the conductor, becoming less as frequency, conductivity or permeability increases.

8.17 Signal line protection

When the I/O line is susceptible to ESD, the line needs to be protected. Usually in case of simple logic devices, diodes are used for protection. They are used for low energy level applications and are not intended for high-energy ones. Such a typical arrangement is as shown in Figure 8.12. Evidently from the figure, it is a simple current limiting arrangement, without any concerns.

Filtering adds another stage of protection by introduction of a filter stage, as shown in Figure 8.13. It should be clear that the filter stage does not modify the data rate, and in case of analog signals, it does not introduce any lags, which can hamper the operation.

Another level of protection is by SPDs, Figure 8.14 shows such an arrangement. One of the major issues while applying surge protection is the length of the lead and/or track. Effectiveness is often reduced by the tracks and/or wire inductance. The length of the track should be as short as possible, and the same holds true for lead lengths. Efforts should be made to run the track as if it is a feed-through device.

The type of filtering used simply depends on the type of device to be protected. Common-mode filtering is one of the different methods used. For instance, wires from a keyboard (i.e., a group of connections) are often subjected to common-mode transients. For such a case, common-mode filtering is the best method; Figure 8.15 shows such an arrangement.

9.1 Introduction

One of the aspects of electromagnetic compatibility (EMC) is that it is difficult to grasp the techniques involved in making measurements. The highlight of this section of engineering measurements can be summarized as follows:

• Measuring in noisy environment
• RF injection, and
• E and H field probes

9.2 Bench and test laboratory

Bench (basic) equipments such as scopes, signal generators, probes, etc can be used for some EMC measurements. These basic equipments are cheap and generally used for relative measurements only. The other are dedicated equipment such as spectrum analysers, antennas, LISN (Line Impedance Stabilizing Network), shielded enclosures, etc that are generally used for troubleshooting and pre-compliance testing. Basic equipments are generally a fraction of the price of dedicated equipments.

Only calibrated equipment such as those found in a test laboratory can perform absolute measurements. Bench equipments are usually used for indication purpose only or for relative measurements, on the other hand, laboratory test equipments are used for pre-compliance and full compliance testing, and are generally costly. Relative measurements can be performed if a benchmark has already been established. For example, a previously measured level A dB is known, after improving the design and by using bench measurements it was found that the level has dropped down by 20 dB. The estimated level now is (A – 20) dB.

Relative measurements can also be used if a sample (golden sample is generally the word used for such a sample) of the product is available with documented test results. By comparing bench measurements of a new sample with the results of the golden sample, estimation can be made of its EMC performance relative to the golden sample.

9.2.1 Work bench ground plane

It is found that common mode noise currents always create problems in measurements. Every instrument alongwith the test sample should be bonded to a metal plane (of Aluminum or Copper). Figure 9.1 shows a typical work bench ground plane. The instruments (used for testing) and prototype should be grounded to the earth plane by short straps rather than using thin wires. Hook up wires and make them run closer to the ground plane so that the noise pick up is reduced. Isolate prototype live wiring from the ground plane (generally spacers are used for this purpose). Use twisted pair of wires for giving DC supply to the prototype.

Using probes

Avoid using long leads and loops (as shown in Figure 9.2) while using any oscilloscope probe measurement in the kHz or MHz frequency range as the loop can cut through the electromagnetic (EM) field and may induce noise voltages, which are not part of the measured unit. This may give an uncertain result that is certainly not expected.

9.3 Determining common mode noise

Figure 9.3 illustrates an excellent way of measuring common mode noise. Short the probe tip and crocodile tip (as shown in the figure) to the point intended for measurement. Ideally, i.e., in a noise free system, it is suppose to measure nothing (that means no voltage). But actually what is seen on the oscilloscope is not what is ideally thought. Actually seen is the noise induced by the common mode currents through the test sample and oscilloscope. This is the prime example of common mode noise being seen by the differential circuit (i.e., the oscilloscopes input).

To reduce the amount of common mode noise that is seen by the oscilloscope use a ferrite across the probe lead and add a few turns to it as shown in Figure 9.4.

Now high frequency common mode currents (that are common to both the positive and negative of the scope lead) see high impedance in the line and therefore reduce the amount of noise flowing through the scope. This works well only into the MHz region, as ferrites do not have high impedance at kHz frequencies.

A much better way of measuring in the presence of high CM noise is to use the differential or a two-probe method as shown in Figure 9.5. Connect one probe tip to the positive end and the other to the negative end of the point intended for measurement. Connect both crocodile clips (i.e., oscilloscope ground) to chassis or plane with respect to which the common mode currents are flowing. Subtract (or invert and add) the two channels for a differential measurement. Always use two identical probes, because if there is any difference in probe impedance then it will cause common mode noise to differential noise conversion and in turn noisy measurements.

9.4 Field probes

9.4.1 Home made near field probes

A near field E-probe can be constructed from a piece of co-ax as shown in the Figure 9.6. The sleeve is pulled back and a monopole antenna is formed. Although this probe can detect near E-fields, it is not directional. A near field H-probe can be constructed with a loop of co-ax. As shown in the Figure 9.6, the inner is connected to the outer braid and a gap in the ends’ outer braid is left at the connection. This probe is prone to capacitive (E-field) coupling so that a ferrite is used to make sure that only magnetic flux is measured. Both of these probes give only approximate indications and can be best used for relative measurements. The commercial versions of both near field E-probes and near field H-probes are also available.

9.4.2 Sniffer probe

A localized sniffer probe should be used when high discrimination is required such as sniffing for high noise traces on a PCB. There are a number of commercial versions available but they can also be constructed in the lab as per the requirement. The probe as shown in Figure 9.7 has got excellent discrimination and can pick up noise from a single component or track. It picks up the magnetic field through the coil and is shielded from other capacitive coupled noise by the tube. The 50 Ω resistor (as shown in the equivalent diagram) matches the source (coil) to a 50 Ω terminated oscilloscope or spectrum analyser.

9.4.3 Home-made current/injection probe

A current probe can be constructed using the home-made H-field probe as shown in Figure 9.8. The current through the wire that is intended for measurement causes the flux to flow through the ferrite and the ferrite concentrates this flux to the centre of the H-field probe, which in turn converts it to a voltage. More windings of either the measured wire or the H-field probe can be used for much higher sensitivity. The fact is that this current probe is only used for rough indication and has a transfer function that depends on –

• Geometry,
• Ferrite, and
• H-field probe

If a signal generator is used instead of an oscilloscope or spectrum analyser, the above set-up can be used as an RF injection probe. Injecting RF currents into individual wires or cable bundles, imitate (i.e., simulate) the effects of these wires being radiated with or being in the presence of high RF fields. This technique (also called BCI, Bulk Current Injection) is more practical than radiation with a RF generator and antenna in a shielded room, and at the same time it is more cost effective.

Figure 9.9 shows the pictures on a CCTV (Closed Circuit TV) monitor showing the effects of RF being injected with the home-made probe. Noise was injected into the positive power line of a bullet camera that is connected to a monitor. Depending on the frequency, different effects will be seen. This simulates the effects of such a system being in the presence of strong RF fields.

9.5 Common mode and Differential mode currents

When using a current probe, care should be taken to distinguish CM (common mode) and DM (differential mode) currents. In a two-wire system such as the supply to equipment, a probe around both wires will measure CM current. To measure DM current, the effect of the CM component can be eliminated as shown in Figure 9.10.

10.2.1 Supply side noise

Any PSU that creates conducted noise can interfere with any other user (load) on the same grid or rail. There are a number of specifications for controlling this problem; these include the IEC-61000-4 range and numerous military, avionics, medical and automotive specifications. These specifications deal with how much of what supply is allowed to generate (back into the power source) and what it must be able to withstand (arriving through/from the power source).

10.2.2 Load side noise

A PSU not only receives power from a source but also supplies a load. Noise generated on the load rails is not as regulated (from a specification point of view) as that of incoming power. This can become a self-compliance problem if the load electronics are sensitive to PSU noise.

10.2.3 AC Supply and PSU generated noise

Figure 10.1 shows the operating frequencies of the AC supply, switch-mode power supplies and microprocessor clock along with their harmonics. Due to 50/60 Hz rectifiers, thyristor switches, load switching etc., utility grid interference usually occurs in the Hz to kHz region. On the contrary, switch-mode power supplies operate/switch in the kHz region with harmonics extending into the MHz region.

10.9.1 Common mode capacitors

Capacitors C_Y1 and C_Y2 attenuate common mode interference and C_X2 is large, have no significant effect on differential mode. The effectiveness of the C_Y capacitors depends very much on the common mode source impendence of the equipment (Figure 10.6). This is usually a function of stray capacitance coupling to earth that depends critically on the mechanical layout of the circuit and the primary-to-secondary capacitance of the mains transformer, and can easily exceed 1000 pF. The attenuation offered by the potential divider effect of C_Y may be more than 15–20dB. The common mode choke is the more effective component, and in cases where C_Y is very severely limited more than one common mode choke may be needed.

10.9.2 Differential mode capacitors

Capacitors C_X1 and C_X2 attenuate differential mode only but can have fairly high values, 0.1 to 0.47 μF being typical. Either may be omitted depending on the detailed performance required, remembering that the source and load impedances may be too low for the capacitor to be useful. For example, a 0.1 μF capacitor has an impendence of about 10 at 150 kHz, and the differential mode source impendence seen by C_X2 may be considerably less than this for a power supply in the hundreds of watts range, so that a C_X2 of this value would have no effect at the lower end of the frequency range where it is most needed.

Figure 10.7 is an example of a mains filter. The earth connection is connected to the metal case.

It seems as if this LC filter is the wrong way around as the noise source (equipment) is on the capacitor side and the load side that should be quiet (mains) is on the inductor side. This is an example of a filter being driven from a high impedance source into a low impedance load.

From a gut feel point of view it can be understood as follows:

• The high frequency noise is shorted through the Y capacitors to earth.
• Any residual noise will see high impedance through the inductor (CM choke) and would rather flow through the capacitor to earth.

11.4.1 Radiated emissions from PCB

In most equipment, the primary sources are currents flowing in circuits (such as clocks), oscillators, etc that are mounted on the PCB. Some of the energy that is directly radiated from the PCB is modeled as a small loop antenna carrying the interference current. Figure 11.3 shown below describes this situation.

A small loop is one that is smaller than a quarter wavelength (λ/4) at the frequency of interest (typical example: 1 meter at 75 MHz). When the loop approaches λ/4 the currents at different points on the loop appear to be out of phase at a distance, so that the effect is to reduce the field strength at any given point. The electric field strength varies with the square of the frequency, and is directly proportional to the signal current and loop area.

E = 263 × 10^-12 × (f² × A × I_s) V/m  …………………..(11.1)

Where, A is the loop area in cm²,

f(MHz) is the frequency of I_s the source current in mA.

From the above equation it is clear that field strength increases with the loop area. The loop area is the path traced by the signal current along with the return path. For the field strength to be as small as possible, the loop area should be small, so one of the rules in PCB design and layout is to keep the area covered by the signal current as small as possible (keep the loop areas small). Keep the signal return path as simple and clear as possible. Avoid puzzling and persistent signal paths as they may cover a large unintended area and that in turn results in increased field strength.

The loop antenna not only acts as a source of emission of unwanted noise but can also receive unwanted noise and in turn is the victim of interference. This is one of the main reasons of using gridded grounds and ground planes, which ensure a defined return path for signal currents and avoids any unintentional return paths.

11.5.1 Radiated emissions from PCBs

Differential mode radiations from small loops on PCBs are not the only contributors to radiated emission; common mode currents flowing on the PCB and on attached cables can contribute much more in comparison. The differential mode currents that are governed by Kirchoff’s current law can be easily predicted, in contrast common mode currents on the PCB are not easy to predict. Figure 11.4 shows the return path for common mode currents via stray capacitance (displacement current) to the other nearby objects.

The full prediction would therefore have to take into account the detailed structure (mechanical) of the PCB and its case, its proximity to ground and to other equipment. The interference current generated in common mode from ground noise developed across the PCB or elsewhere in the equipment and may flow along the conductors or along the shield of the shielded cable. Here even if the cables are longer than λ/20, they will act as an antenna. The common mode noise generally couple through parasitic capacitances.

The equation of field strength in case of common mode is as follows:

E = 1.26 × 10^-4 × (f × L ×I_CM) V/m  …………………..(11.2)

Where, L is the cable length in meters,

I_CM is the common mode current at f MHz in mA flowing in the cable.

Here if the length of the cable increases, the field strength increases proportionally. In PCB design and layout, steps should be taken to reduce common mode currents (minimize common mode currents). Partitioning, filtering, grounding and using planes in PCB design can achieve this.

11.6.1 Signal Integrity Vs EMC

There is a high degree of awareness among digital designers regarding the problems using high speed signals on PCBs. The problems such as ringing, mismatch and cross-talk on PCBs can cause upsets in digital logic. The general concerns to Signal Integrity (SI) are:

• Power Decoupling
• Ground Bounce
• Reflections
• Cross Talk

A lot of effort is made to design circuits having signal integrity i.e. to be self compliant and extension of this effort will bring us to total EM compatibility. Implementing the best EMC design practices and techniques available will help us achieve Signal Integrity; good signal integrity will make the board more immune to interferences and in turn quieter. The techniques for PCB design are dealt with in the following sections and are mentioned as below:

• Board layout
• Planes and layering
• Grounding
• Decoupling
• Transmission lines
• Multiple boards

12.2.1 Malfunction of systems

Digital bus architecture in which many signals are multiplexed on a single bus under software control (as in processor based systems) is preferred over hard-wired logic (as in relay logic). Such a structure (digital) is more prone to interference because a low level of energy is sufficient to induce change of state. A random pulse may or may not affect the operation because it depends on its timing with respect the clock, data transmitted and a program’s execution state, it is impossible to predict the effects of the interference. If the interference exceeds the tolerable limit (in this case logic threshold), the processor operation will be disrupted which leads to system failures.

Phenomena

Electromagnetic phenomena expected to interfere with systems are given below:

• Power supply dips, surges, fluctuations, interruptions;
• Transients on power supply lines, control lines and signal lines;
• Electrostatic discharge (ESD) from a charged object;
• Low frequency magnetic and/or electric field;
• Radio frequency fields coupled within the equipment or through connected cables.

Software

Many times malfunctions due to software are confused with those due to EMI. In real time systems, critical software execution along with transients of external conditions causes operational failures that lead to system crashes, incorrect data and faulty operation – identical to those induced by EMI. This makes it difficult to distinguish faulty software from poor EMC practices.

Interference with radio reception

In the radio spectrum, if the interference is present on some wanted channel then that channel will be destroyed if the interference is of same or greater magnitude. Equipments should be immune from the local fields of intentional radio transmitters and the unintentional emission must be limited to protect the operation of intentional radio receivers; from this it is clear that the complementary aspects of EMC (emission and immunity) address two different issues.

Certain industrial, scientific and medical equipments (collectively known as ISM equipments) generate high level of RF energy that is not used for communication and also produces interference. Co-channel interference is caused when two radio systems are allowed to use the same frequency on the basis that there is sufficient distance between the systems. But due to abnormal propagation, interference occurs that is often regarded as an EMC issue.

Disturbance on mains supply

Disturbances caused by voltage variations, fluctuations and interruptions, waveform distortion, transients and surges leading to malfunction in systems are referred to as EMC issues.

The mains distribution network like main signaling (MS) superimposes signals in the frequency range of 3 kHz to 150 kHz on the mains which is the also the best efficiency operating region for electronic power converters like switch-mode power supply (SMPS), variable speed motor drives, induction heaters, etc. This supply in turn is used by the supply industry itself and also by the consumers.

Other EMC issues

All the above-mentioned issues directly affect the product design but there are some other issues that are necessary to be known. They are as follows:

EEDs (electro-explosive devices) and flammable atmospheres

The incident RF energy results in the ignition of flammable atmospheres in petrochemical plant while EEDs are connected to their power sources for detonation, which can behave as antenna.

Data security

Security of confidential data is another aspect of EMC. Low RF emission from the data processing equipments may be modulated with the information. These signals can be detected by the third party and can demodulate for their own purpose.

Electromagnetic weapons

An intentional broadband radiated pulse can be generated and it can upset all the susceptible electronics within a certain range. This can be used as a weapon against enemies.

13.1 Introduction

It is known that EMC has two complementary aspects of emissions and immunity. But the roots of these two aspects are altogether different.

Control of emissions

In most countries of the world, the radio spectrum is extensively used for many kinds of traffic applications. Broadcasting and telecommunications are the most obvious and common uses, and telemetry, radar, radio navigation and space research are some of the other uses. Spectrum users pay a license for the right of being allowed to transmit and receive in a particular spectrum and in return they expect that this right should not be affected by interfering sources. In addition to this, the safety related services demand assured reliability of spectrum access.

For these reasons governments have found it necessary to regulate the spread of types of apparatus that, though not licensed as radio transmitters (i.e., unintentional radiators/transmitters) have the potential to interrupt such services. Historically, such apparatus has been dominated by broadband sources such as motor-driven equipment, fluorescent lights and pulsed ignition from petrol engines.

Electronic equipments also have the potential to generate radio frequency interference (RFI) as a by-product of its operation, until and unless it is intentionally designed to avoid doing so. The main culprits are digital electronics incorporating microprocessors, and power switching circuits using fast electronic switches such as MOSFETs, IGBTs, transistors, TRIACs, etc. Even though they are not intended to transmit RF energy, sufficient energy is produced in the sensitive parts of the spectrum to make it necessary to apply the same types of legislative control as has been applied to other types of unintended source in the past.

With the extension to the concept of EMC to include compatibility with the supply network, control of other types of emissions is becoming mandatory. These are primarily power frequency harmonic currents and flicker. Because the mains power supply at a given node may be shared among several users and has finite source impedance, any load current disturbances belonging to one user will cause voltage distortion that is introduced to all the other users at that point. The amplitude of such distortion has to be limited, and since the source impedance cannot economically be made arbitrarily low, this suggests some limit on the allowable current disturbances.

Control of immunity

The immunity aspect of EMC is more complex and quarrelsome issue such that it has several strands that need to be separated and examined:

• the need for continuous safe operation/functioning of safety related systems in the face of external interference
• the idea of economic tradeoff in sharing requirements as between emissions and immunity
• the question about fitness for the purpose of apparatus intended for a given environment
• the wish to establish a baseline for product quality, and hence discard the consumer’s option to pay less (that to for less reliable products).

Safety aspects

For any system that could present a hazard to people (or property) if it was not correctly controlled, safety questions must invariably be addressed, and these questions must include the issue of safety in the face of external interference. Because the inherent susceptibility to electromagnetic disturbances of electronic control devices has been understood for many years, it is customary for true safety-critical functions to be controlled only by electro-mechanical means. For example, the emergency stop on a large machine will be simply a switch that physically disconnects the power to the machine when it is activated. On the other hand, on more complex systems, relay logic might be used for the same function. The greatest advantage of such an approach is that electromechanical devices are immune to normal electromagnetic disturbances and they simply don’t have enough energy to cause a change of state. However, the more complex the control systems become, the harder it becomes to implement safety critical functions in such a simple form, and the question of immunity of the electronic control devices has to be addressed.

One day, no product will be sold without being EMC compliant. The need for EMC Engineering will always be there. Figure 13.1 shows the Electromagnetic Interference (EMI) within systems, i.e., Inter-system EMI.

13.2 EMC Legalities

There are few of the points that are to be made clear before making EMC legal. One of them is the Compliance; it should be made mandatory and at the same time has to be ensured. Next comes the Standards that are to be applied and definitely the Cost has to be well under control. There are a number of factors that really does matter with incorporating EMC. Better EMC results in –

• improved reliability
• better signal integrity
• competitive advantage
• fewer product returns
• lower life cycle cost
• nice healthy feeling

Tutorial 1: Common Mode Impedance Coupling & PCB Track Size

There are two important parameters when designing power connections, whether it is a PCB track or type of wire. One is the voltage drop across that connection/track/wire and the other is its current carrying capability. Both depend on the connection resistance/impedance.

If the impedance/resistance is too high, too much voltage will drop. The higher the R, the more energy the connection will dissipate. This can ultimately lead to the track/wire burning out.

This tutorial also shows why it is better to use a single point (parallel) or star point ground.

Part A

Solution

The track can be seen as a resistance or common impedance in the ground of the two circuits. Calculate the resistance of the copper in the track and then the voltage drop across it. This will be the ground bounce in the analog circuit with respect to any off-board ground.

The ground jump (noise voltage) peaks are almost 0.5V !

Part B

Solution

Use the graphs in MIL-STD-275E (or the latest version document no.: IPC-D-275 at www.ipc.org) and determine the temperature of the track above ambient. Use the rms value of current for this

According to the graph for a track that is not embedded it will rise with 30 to 40deg C. This is high. A figure of 10deg or max 20deg is preferable.

For 5A to only cause 10 degC rise one would need a track cross section of about 150 sq. mils.

This means a track width of:

Use a 3 mm track width.

The voltage drop will now be:

Area: =0.003m·35·10^-6m Cross section area of a 1oz (35 micron) copper strip 3mm wide

The analog circuit will see voltage peaks of: V:= R·10A V = 0.164V

This voltage drop may be more acceptable.

The best solution though is to use two different tracks.

One for the power circuit and one for the analog circuit. A single point parallel or star point ground. The 0.16V drop in the power circuit might still not be acceptable for that circuit, but at least the analog ground is clean!

Tutorial 2: Emissions and Solutions

The solution

Is it possible for the circuit to emit at 100MHz?

Yes. It was established that the logic used has rise/fall times of less than 1nsec. The associated bandwidth of the noise is:

The capacitance between the PCB and enclosure is therefore:

The 20MHz clock and its harmonics also emit, but the logic edges have the higher spectrum.

Why did taking off the lid not make a significant difference?

The answer lies with the difference between common mode and differential mode emissions.

Differential mode emissions are from the loop areas on the PCB. Opening the lid let the DM emissions “out”. This is relatively small compared to emissions from the long cable sticking out of the enclosure.

Assume that the largest loop is 1cm by 1cm and that the total current flowing in these loops is 5mA. At a distance 10m away from the circuit:

13μV/m is relatively low emissions. Compare this to a typical 2mV/m that is required for good 100MHz VHF reception.

If of course the problem of the long radiating power lines is solved, the lid might become an issue if you really want absolute no interference.

Should the leakage trough the lid be an issue, keep the screws apart with not more than:

This is the lambda over 20 rule for apertures. In this case screws 150mm apart would degrade the enclosure screening with 20dB. So keeping the screw distances (or any aperture/hole sizes for that matter) less than say 50 – 100mm is OK.

Note that the area used here is smaller than that of the physical PCB, as it will only be the area under one gate. All the other gates probably cancel each other out, as if a logic state goes high and another low, their effects cancel out

Why is the power cable emitting?

Common mode currents are generated by the switching logic. To get a feel for this consider the following:

The board and the bottom of the enclosure is acting like a capacitor. This capacitance can be calculated by knowing the area and separation h.

The above shows that there is around 0.6pF of capacitance between the PCB and enclosure.

Approximate the inductance of the wire by using 1.5nH per mm. Twice the lenght of the distance l is used (there and back)

The common mode loop will have an impedance of say:

A typical common mode current will flow:

At a distance of 10m, the CM emissions will be:

This is much higher than the Differential Mode Emissions.

Why did the cable shielding not work?

Grounding the cable at only one end is typically done for analog and audio noise below 1MHz. At 100MHz and if not grounded at both ends, the shield will act like a monopole antenna.

When the shield was grounded at both ends, there were long pigtails. The pigtails (at about 1.5nH per mm) will resonate with any capacitance. This includes PCB to enclosure capacitance and stray capacitance between the cable and earth. If a resonance of this LC circuit is around 100MHz, the cable will emit even better than before.

What is the practical solution?

Three solutions. Use them on their own. Combinations will work better though.

1. Use a complete shielded cable. Terminate the shield 360( with proper back shells on the connectors. If all RF current carrying circuits and connections are tightly screened/shielded, the emissions will be reduced. Depending on the frequency – good screening can give you a good 60dB and more.
2. Use capacitors to chassis or other types of feed through filters at the entry point. This will dump or short the CM current to chassis (enclosure). There are different types and sizes available. 40 to 60dB of noise current reduction is possible.
3. This is an easy one. Soft ferrites work very well at 100MHz. You even get them in the ‘clip-on’ variety. At 100MHz a small ferrite bead can add a few hundred ohms of resistance to the CM noise current. If you wind the wire through the bead a few times, this can go up to kilo-ohms.

IDC Technologies EMC “SG” Practical Kit

2. Radio in a Box

Place tuned FM radio with antenna in box. Close lid. No reception. Draw antenna wire from hole – reception resumes. This illustrates how emissions will take place from enclosure with UNFILTERED conductors/cables. Figure 1.

Use feed-trough filter to attenuate reception. Ground the bulkhead/feedtrough filter directly to the enclosure chassis. This illustrates the use of feedtrough filters when feeding cables trough a shielded enclosure/faraday cage boundary. Figure 2

Use the earth wire to ground the filter. The filter efficiency reduces (does not work). This illustrates that there is too much impedance for FM reception (around 100MHz) in a round wire. It also illustrates how filters can be ineffective – i.e. bad grounding. At 100MHz a round wire cannot be used as bonding/grounding. Figure 3.

With the box closed, feed the antennal wire through the cable shield/braid. If the braid is not grounded to the enclosure, nothing happens. Ground the shield to the open connective part of the box. Reception is attenuated. Figure 4.

Use the round “earth” wire to ground the shield. This does not work and illustrates that the wire is not a good conductor at FM frequencies. It also illustrates why pigtails on cable screens diminishes the shielding effectiveness – Figure 5.

NOTE:

Before performing demonstrations – ensure batteries for the radio (2 X 1.5V penlight) and PSU (2 X 9V ‘522’) are ‘fresh’.

3. Cell Phone Shielding

Enclose a cell phone (not included) inside the box. Tighten the screws. Contact the mobile phone. If properly shielded it should not receive the call. Note the blue EMC gasket in the lid. If it still receives, plug the large hole with metallic gauze (or steel wool). Figure 6.

Feed a wire through one of the smaller holes. Depending on signal strength, the phone should now receive an incoming call.

4. Common Mode Choke

Use the toroid and wires to illustrate the windings and working of a CM choke.

5. PSU & Filter

Use the jumpers on the PSU filter to show effects of different filter elements. Red WIMA is C_x. Two black C’s are Y-caps. Use the ground extension strap to connect PSU plane to scope. This will accentuate CM effects. Jumpers closest to input selects CM choke. Figure 8.

6. Microwave Oven Safety Meter

Use a cell phone (operating around 800 – 900MHz) to transmit and measure at the antenna with the microwave oven meter (Figure 9). Through the back of the meter (Figure 10) a diode/crystal can be seen connected directly to the meter. This illustrates demodulation of RF by a diode.

8. Faraday Shield and Ferrites

Use a bag of copper mesh and a small portable radio.

Replace the antenna with a piece of wire. The wire is insulated. If the antenna makes contact with the screen it doesn’t work.

If the FM radio is dropped in the bag, it stops receiving.

It works with cell/mobile phones as well, but the cell phone is harder to silence (because of higher frequency).

If the antenna wire is taken through a hole in the mesh (shield), the radio starts working. This shows that if any conductor is passed (unfiltered) through a shield, the shielding is violated. Works with well for FM.

Use an EMI suppressing ferrite and wrap a few turns around it. If outside of the shield – reception is hampered. If the radio is in the shield but the ferrite filters the antenna breaking through the shield, it doesn’t receive – as opposed to the situation without it. Shows how good the ferrite works!

The ferrite does not have the same effect on AM reception < 2MHz. Ferrites work well around 100MHz!

9. DC-DC Converter and Common Mode Noise

Use a switch mode DC to DC converter. A small rechargeable lead-acid battery is a good supply as it will not generate any noise – as opposed to a bench supply. Load the DC-DC with resistors. A number of effects can be demonstrated:

1. Use an oscilloscope and show switching noise on the input/supply side of the DC- DC. Scope on AC and a few mV resolution.
2. Use an AM and/or FM radio to pick up switching noise from the supply.
3. Show that the radiated noise comes from the supply side wires. The conducted noise that now emits is on the cables all the way to the supply/battery/bench supply and is not just localised around the DC-DC itself.
4. Connect 1nF to say 10nF caps to all the inputs and outputs and bring the other side of the caps together. In a design these caps will be connected to chassis. Show that when the common point of the caps are connected to chassis (typically metal enclosure) of the DC-DC, the noise dies down. Illustrate with either the radio and/or the scope. The ‘chassis’ caps dump or short circuit common mode noise from the inputs/outputs to the chassis so that it does not travel along the cables.

5. Connect the DC-DC input through a common mode choke. Again illustrate that the noise is reduced. A CM choke with ‘chassis’ caps on the input of the DC-DC is a very good CM filter. If the effect is not dramatic enough, use more windings and/or a larger choke.
6. Since DC-DC noise is usually low, clip-on ferrites will probably not work so well but is worth a try.

Simulating Radiated Susceptibility by Bulk Current Injection

In this demonstration a home made current probe is used in reverse to inject RF onto cables.

This technique is used when it is not practical or too expensive to subject a complete system to RF illumination. Good example is an aircraft. Instead of parking a jumbo jet in a test facility chamber, RF is injected onto the cable bundles to test immunity of on-board equipment. This simulates the cables and wires acting as antennas and carrying emissions into equipment.

Construct a home made current/injection probe. Use a clip-on ferrite. Construct a loop antenna (magnetic near field probe) by connecting the inner conductor of a co-ax to the shield. Leave a small gap in the shield where the inner conductor was used.

This can either be used as an injection probe or current measuring probe.

For this demonstration use the probe with a signal generator and inject noise into various cables of equipment.

The following show the noise induced on a surveillance CCTV’s power leads using a signal generator around 30MHz (30Vp-p) and the home made injection probe. The two pictures are for the normal signal and noise induced signal.